Magnetic field response measurement acquisition system

ABSTRACT

Magnetic field response sensors designed as passive inductor-capacitor circuits produce magnetic field responses whose harmonic frequencies correspond to states of physical properties for which the sensors measure. Power to the sensing element is acquired using Faraday induction. A radio frequency antenna produces the time varying magnetic field used for powering the sensor, as well as receiving the magnetic field response of the sensor. An interrogation architecture for discerning changes in sensor&#39;s response frequency, resistance and amplitude is integral to the method thus enabling a variety of measurements. Multiple sensors can be interrogated using this method, thus eliminating the need to have a data acquisition channel dedicated to each sensor. The method does not require the sensors to be in proximity to any form of acquisition hardware. A vast array of sensors can be used as interchangeable parts in an overall sensing system.

CLAIM OF BENEFIT OF PROVISIONAL APPLICATION

Pursuant to 35 U.S.C. § 119, the benefit of priority from provisionalapplications having U.S. Ser. Nos. 60/467,844, filed on Apr. 30, 2004;60/467,840, filed on May 1, 2003; 60/467,841, filed on May 1, 2003;60/467,113, filed on May 1, 2003; 60/467,839, filed on May 1, 2003; and60/467,842 filed on May 1, 2003; 60/467,112, filed on May 1, 2003; and60/467,194, filed May 1, 2003 is claimed for this nonprovisionalapplication.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is related to co-pending, commonly owned patentapplication Ser. No. ______, filed Apr. 30, 2004, entitled “MagneticField Response Sensor for Conductive Media.”

STATEMENT REGARDING FEDERALLY SPONSORED RESEARCH OR DEVELOPMENT

The invention described herein was made in part by employees of theUnited States Government and may be manufactured and used by and for theGovernment of the United States for governmental purposes without thepayment of any royalties thereon or therefore.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates generally to a remote monitoring system.It relates in particular to a monitoring system comprising one or moresensors, which utilize L-C (inductance-capacitance) or L-C-R(inductance-capacitance-resistance) resonant circuits, in combinationwith an interrogation means, to monitor a variety of properties,including strain, temperature, pressure, identification, performance,chemical phase transition (such as melting and state-of-cure), fluidlevel, wear, rotation rate, location and proximity. The systemeliminates the need for physical connection to a power source (i.e., nolead wires) or to data acquisition equipment, and allows for multiplemeasurements using a single acquisition channel. Additionally, it doesnot require that the sensors be in proximity to any form of acquisitionhardware and it facilitates use of a portable handheld interrogationunit.

2. Description of the Related Art

A magnetic field response sensor is a passive inductor-capacitivecircuit designed to change correspondingly with a change in the physicalstate that the sensor measures. Use of inductors and capacitors to formresonant circuits is established in the literature. See, for example, D.Halliday and R. Resnick, Fundamental of Physics, 2nd Edition, Wiley, NewYork, pp. 624-634 or similar basic physics or electronics texts.Wireless measurement acquisition systems that use existing sensorsphysically connected to a power source, microprocessor and transmittersare described in Woodard, S. E., Coffey, N. C., Gonzalez, G. A., Taylor,B. D., Brett, R. R., Woodman, K. L., Weathered, B. W. and Rollins, C.H., “Development and Flight Testing of an Adaptable VehicleHealth-Monitoring Architecture,” Journal of Aircraft, Vol 40, No. 5,September-October 2003. A method of acquiring measurements without theneed for physical connection to a power source is the use of radiofrequency identification (RFID) tags. This method relies on the use ofradio-frequency integrated circuits functionally coupled to sensors.Representative of patents covering RFID tags is U.S. Pat. No. 5,420,757.An example of a system for interrogating fluid level is that presentedby Kochin et al. in U.S. Pat. No. 6,335,690, which teaches a preferredseparation distance between the sensor and the interrogator of less than3.5 cm. U.S. Pat. No. 6,111,520 (Allen) and Fonseca, M. A., English, J.M., Arx, M. V. Allen, M. G., “High Temperature Characterization ofCeramic Pressure Sensors,” Proceeding of 1999 IEEE MEMS Workshop, pp146-149 discuss several methods of magnetic field response sensorinterrogation having the sensors within the perimeter of the antennaused for interrogation. Planar or laminar designs of L-C circuitsinclude rectangular inductors (e.g., U.S. Pat. No. 6,025,735), spiralinductors (e.g., U.S. Pat. No. 6,111,520), parallel place capacitors(e.g. U.S. Pat. No. 6,335,690) and interdigitated capacitors (e.g., seeK. G. Ong and C. A. Gaines, Smart Materials Structure, (9) 2000;421-428).

Key to the practical use of a measurement acquisition system, increasedantenna-sensor separation distance, multiple measurements whose dynamiccharacteristics affect different attributes of the sensor's magneticfield response, and portability of the interrogation unit is desired.

SUMMARY OF THE INVENTION

Accordingly, it is an object of the present invention to provide amagnetic field response measurement acquisition system having increasedinterrogation antenna and sensor separation distance.

Another object is the interrogation of multiple sensors concurrentlyusing a single acquisition channel.

Another object is to provide a magnetic field response measurementacquisition system having a portable interrogator.

An additional object is to provide a magnetic field response measurementacquisition system enabling the easy incorporation of additionalsensors.

Another object is to provide a magnetic field response measurementacquisition system capable of acquiring more than one measurement fromeach sensing element.

A further object is to facilitate multiple measurements whose dynamiccharacteristics affect different attributes of the sensor's magneticfield response.

Additional objects and advantages of the present invention are apparentfrom the drawings and specification which follow.

In accordance with the present invention, a magnetic field responsewireless measurement acquisition system comprises an interrogator whichmay be portable and handheld, at least one inductively powered L-Csensor, and software to determine sensor properties (e.g., resonantfrequency, bandwidth, amplitude, etc.). The interrogator and softwarecan be used with L-C sensors that measure a variety of parameters,including temperature, pressure, strain, location, rotation rate, andother parameters. The sensors convey basic waveform information (e.g.,frequency, bandwidth, etc.) that is dependent solely on the propertiesbeing measured, and do not require wide bandwidths to transmit modulatedinformation. The sensors emit a single radio frequency (RF)transmission, thus there is no requirement that information betransmitted as a modulated signal on the RF carrier. As a result, thesensors can be designed to have a higher Q (i.e., narrower bandwidth)than existing wireless sensing systems. This higher Q sensor can beinterrogated at a greater distance and at lower power than lower Qsensors. There is also potentially less interference from neighboringsensors and higher sensor densities. Additionally, simplified systemarchitecture enables the interrogator to be built into a handheld unit.An algorithm quickly determines the characteristic sensor parameters inan efficient manner, not requiring storage of readings across a spectralrange and subsequent analysis of the ordered pairs. A vast array ofsensors can be used as interchangeable parts in an overall L-C sensingsystem.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic of an embodiment of an L-C measurement acquisitionsystem in accordance with the present invention.

FIG. 2 is a schematic of magnetic field response sensor measurementbands.

FIG. 3 is a flowchart illustrating interrogation logic.

FIG. 4 is a graph of sensor response amplitude as excitation frequencyapproaches sensor resonant frequency.

FIG. 5 illustrates resistive response curves.

FIG. 6 is a schematic of the interrogation system.

FIG. 7 illustrates a sensor circuit.

FIG. 8 is a representative antenna.

FIGS. 9 a and 9 b are graphs of resistance measurements.

FIGS. 10 a and 10 b are graphs of inductance measurements.

FIGS. 11 a and 11 b are graphs of quality factor, Q.

FIG. 12 illustrates a square spiral inductor.

FIG. 13 is a graph of resistance versus inductor trace width.

FIG. 14 is a graph of quality factor, Q, versus inductor trace width.

FIG. 15 is an illustration of a sensor mounted on a conductive surfacevia a spacer.

FIG. 16 is an illustration of a sensor mounted on a conductive surfacewith the inductor projected away from the conductive surface.

FIG. 17 is a schematic of a conductive closed cavity sensorconfiguration.

FIG. 18 is a schematic of a sensor for a conductive closed cavity.

FIG. 19 is a schematic of a conductive cavity with antenna and multiplesensors located internal to the cavity.

FIG. 20 a is a schematic of a sensor embodiment for phase transition andstrain measurement.

FIG. 20 b illustrates a sensor embodiment that can be used todistinguish parts during curing

FIG. 21 is a graph of time history of sensor response during resincuring.

FIG. 22 is a schematic of a sensor embodiment for wear or thermalmeasuring utilizing interdigital electrodes.

FIG. 23 illustrates an embodiment of an interdigital device with one ofthe electrodes having a temperature sensitive dielectric or a dielectricwhich has a phase transition when exposed to excessive temperature

FIG. 24 illustrates a sensor embodiment for wear or thermal measurementhaving the inductor embedded within the capacitor.

FIG. 25 illustrates a sensor embodiment for wear or thermal measurementhaving the inductor mounted externally.

FIG. 26 a sensor embodiment for wear or thermal measurement having asensor embedded in a cube.

FIG. 27 illustrates interdigital electroplates.

FIG. 28 illustrates an embodiment of interdigital electroplates withtemperature sensitive dielectric, thermomagnetic or a phase transitiondielectric between the electroplates.

FIG. 29 illustrates a capacitor with a negative electroplate thattranslates perpendicular to its surface and a stationary plate.

FIG. 30 illustrates an embodiment of a sensor for displacementmeasurements.

FIG. 31 is a first graph of capacitor variation with displacement.

FIG. 32 is a second graph of capacitor variation with displacement.

FIG. 33 illustrates a second embodiment of a sensor for displacementmeasurements.

FIG. 34 illustrates a third embodiment of a sensor for displacementmeasurements.

FIG. 35 is a graph showing capacitance variation with displacement.

FIG. 36 illustrates a fourth embodiment of a sensor for displacementmeasurements.

FIG. 37 illustrates electroplates and dielectric medium for a firstembodiment of a sensor for fluid level measurements.

FIG. 38 illustrates a first embodiment of a sensor for fluid levelmeasurements.

FIG. 39 illustrates electroplates having residual fluid film.

FIG. 40 illustrates n pair of parallel electroplates and dielectricmedium for a second embodiment of a sensor for fluid level measurements.

FIG. 41 illustrates a second embodiment of a sensor for fluid levelmeasurements.

FIG. 42 illustrates a third embodiment of a sensor for fluid levelmeasurements.

FIG. 43 illustrates a cross section of interdigital capacitor withelectric field.

FIG. 44 illustrates a dielectric medium in contact with electrode pairs.

FIG. 45 illustrates electrodes with a thin residual film.

FIG. 46 is a graph of frequency measurement for two fluid level sensors.

DETAILED DESCRIPTION OF THE INVENTION

Referring now to the drawings, and more particularly to FIG. 1, anembodiment of a magnetic field response measurement acquisition systemin accordance with the present invention is shown and referencedgenerally by numeral 10. Acquisition system 10 will first be describedin terms of a general overview with the aid of FIG. 1.

Radio Frequency (RF) broadband antenna 12 transmits and receives RFenergy. Processor 14 regulates the RF transmission and reception.Processor 14 includes algorithms embodied in software for controllingthe antenna 12 and for analyzing the RF signals received from the one ormore magnetic field response sensors 16. Sensors 16 are passiveinductor-capacitor L-C circuits or inductor-capacitor-resistor L-C-Rcircuits. Each inductor L is placed in parallel with a capacitor C,forming an L-C(p) circuit. Processor 14 modulates the input signal tothe antenna 12 to produce either a broadband time-varying magnetic fieldor a single harmonic magnetic field. The variable magnetic field createsan electrical current in the sensors 16 as a result of Faradayinduction. Each sensor 16 will electrically oscillate at resonantelectrical frequencies that are dependent upon the capacitance andinductance of each sensor 16. The oscillation occurs as the energy isharmonically transferred between the inductor (as magnetic energy) andcapacitor (as electrical energy). When the energy is in the inductors,the magnetic fields produced are single harmonic radio frequencies whosefrequencies are the respective sensor 16 resonant frequencies, and aredependent on how the physical measured property changes the capacitanceof the circuit. The antenna 12 is also used to receive the harmonicmagnetic responses produced by the inductors. The receiving antenna canbe the same antenna used to produce the initial broadcast of energyreceived by the L-C circuit or another antenna can be used. When thesame antenna is used, it must be switched from a transmitting antenna toa receiving antenna. A simple microprocessor can be used to identify thefrequencies of the signals received by the antenna 12. The measuredfrequencies are then correlated to measurement of physical states.

As illustrated in FIG. 2, the sensors 16 are designed such that theirrange of measurement frequencies do not overlap, but are within afrequency range of the antenna 12. The individual ranges of resonantfrequencies correspond to physical property values that can be measured.The capacitors are designed such that, when electrically coupled to theinductors, their range of values will be a predetermined partition ofthe RF frequency band. This method allows for any number of sensors 16within the range of the antenna 12 to be interrogated concurrently.

The use of magnetic field sensors 16 and the measurement architecture ofthe present invention greatly reduces measurement acquisitioncomplexity. The magnetic field response sensor 16 is a passiveinductor-capacitive circuit designed to change correspondingly with achange in the physical state that the sensor 16 measures, and acquirespower via Faraday induction. Sensing is provided by measuring resonantfrequency shifts due to changes in inductance or capacitance, requiringno batteries. The harmonic magnetic field response of the inductorserves as a means of transmitting the resonant. Key attributes of themagnetic field response are amplitude, frequency and bandwidth. Thesensors 16 can be designed such that one of the attributes variescorrespondingly with the measured physical state. A RF antenna canproduce the time varying magnetic field used for the Faraday induction,as well as receive the magnetic fields of the the sensor 16. The use ofmagnetic fields for powering the sensors 16 and for acquiring themeasurements from the sensors 16 eliminates the need for physicalconnection from the sensor 16 to a power source and data acquisitionequipment. The architecture also eliminates the need to have a dataacquisition channel dedicated to each sensor 16. Multiple concurrentmeasurements can be accomplished with a single acquisition channel andmultiple sensors, each with a different resonant frequency, can beprobed by the broadband antenna 12.

Capacitor geometric, capacitor dielectric, inductor geometric orinductor permeability changes of a sensor will result in magnetic fieldresponse frequency change. Any resistive change will result in aresponse bandwidth change. Dielectric variations (e.g., due to thepresence of chemical species or due to a material phase transition) tothe capacitor can be designed for specific measurements. Further, aresistive element whose resistance changes with a physical parameter canalso be placed in circuits of fixed capacitance and inductance. Hence,the system has the potential for acquiring many different types ofmeasurements. Because the sensors' 16 functionality is based uponmagnetic fields, they have potential use at cryogenic temperatures,extremely hot temperatures, harsh chemical environments and radiativeenvironments.

When a sensor's 16 inductor comes in proximity to a conductive material,energy is lost in the sensor due to eddy currents being produced in theconductive material. As the sensor is brought closer to the material,the response amplitude decreases while the response frequency increases.Hence, this effect can be used to determine proximity to conductivesurfaces. Otherwise, it is necessary to maintain a fixed separation. Ifcapacitance and inductance are fixed, changes to a sensor's 16orientation or position with respect to interrogating antenna 12 changesresponse amplitude. The interrogation system of the present inventionallows for the acquisition of measurements from any magnetic fieldresponse sensor 16 developed to exploit the aforementioned phenomena.The system also allows for autonomous sensor interrogation, analysis ofcollected response to value of physical state and comparison of currentmeasurements with prior measurements to produce dynamic measurements.

The measurement acquisition method can be used to acquire measurementseven when the sensor 16 is embedded in material that is transmissive tothe RF energy that interrogates the sensor 16. An advantage of thismethod is that the components for the method can be non-obtrusivelyadded to the vehicle/system for which it is being used. An antenna 12can be produced as a metallic foil or as metal deposited on a thindielectric film. Either aforementioned version of the antenna 12 can bemounted to an existing bulkhead or other structural component. For someapplications, sensors 16 can be fabricated using metal depositionmethods. Metal deposition can be used to add sensors to avehicle/structure during manufacturing. Other advantages of the methodinclude (1) no line of sight being required between the antenna 12 andsensor 16, (2) the ability of the entire sensor 16 to be embedded in anonconductive material, (3) the ability to embed the capacitive elementin a conducting material with the inductive element being placed awayfrom the surface of the conductive material, (4) no specific orientationof the sensor 16 with respect to the antenna 12 is required except thatthey cannot be 90 degrees to one other, and (5) no wiring is required toadd new measurements, only a partition of a RF bandwidth used in themeasurement spectrum and a frequency/measurement correlation table.

Interrogation

Interrogation utilizes a scan-listen-compare technique, which allows forhigh signal-to-noise ratio. FIG. 3 illustrates the interrogation logic.Separate transmission and receiving antennae can be used or a singleswitching antenna can be used. Using two antennae provides a largervolumetric swath at which measurements can be taken, which isapproximately double that of a single antenna. The interrogationprocedure generally comprises the following steps:

-   -   (a) At the lower limit of a predetermined range, a radio        frequency harmonic is transmitted for a predetermined length of        time and then the transmission mode is swtiched off (i.e., the        transmission antenna is turned off if two antennae are used or,        if a single antenna is used, it ceases transmission).    -   (b) The receiving mode is then turned on (i.e., the receiving        antenna is turned on if two antennae are used or, if a single        antenna is used, it begins receiving). The received response        from the sensor 16 is rectified to determine its amplitude. The        amplitude, A_(i)(t), and frequency, ω_(i)(t), are stored in        memory.    -   (c) The receiving mode is turned off and the transmission mode        is turned on. The transmitted radio frequency harmonic is then        shifted by a predetermined amount. The harmonic is transmitted        for a predetermined length of time and then the transmission        mode is turned off.    -   (d) The receiving mode is turned on. The received response from        the sensor 16 is rectified to determine its amplitude. The        amplitude, A_(i), and frequency, 107 _(i), are stored in memory.    -   (e) The current amplitude, A_(i), is compared to the two        previously attained (recorded) amplitudes, A_(i-1) and A_(i-2).        If the previous amplitude, A_(i-1), is greater than the current        amplitude, A_(i), and the previous amplitude A_(i-1) is greater        than amplitude prior to it, A_(i-2), the previous amplitude,        A_(i-1), is the amplitude inflection. The amplitude inflection        occurs when the excitation harmonic is equal to the resonant        frequency of the sensor 16. The amplitude, A_(i-1), and the        corresponding frequency, ω_(i-1), are stored for the sensor 16        for the current frequency sweep. These values can be compared to        the values aquired during the next sweep. If an amplitude        inflection has not been identified, then steps (c) and (d) are        repeated.    -   (f) If amplitude inflection has been identified, the harmonic        sweep continues to the next sensor 16.

FIG. 2 illustrates three antenna sweeps for n sensors 16. The initialfrequency sweep can be used to identify and catalog (store) all keyresponse attributes (resonant amplitudes and frequencies) associatedwith all n sensors 16 within the antenna's 12 range of interrogation. Ifa particular sensor 16 is resistive, its bandwidth will also be stored.The cataloged resonant amplitudes and frequencies for all sensors 16 canbe used to reduce the sweep time for successive sweeps. For example, thenext sweep to update each resonant frequency can start and end at apredetermined proximity to the cataloged resonant and then skip to thenext resonant. Every sensor 16 does not need to be interrogated duringeach successive sweep. The interrogation rate for each sensor 16 shouldbe dependent upon the rate that the physical state that sensor 16measures changes. FIG. 2 illustrates interrogation of sensor 21 andsensor n during the second sweep. Sensors 21, 22 and 23 have frequency,bandwidth or amplitude changes corresponding to variations in theirmeasured physical states. Sensor n only has amplitude variationscorresponding to either a displacement or rotation measurement.

Measurement resolution is also depicted in FIG. 2. Each sensor 21, 22,23 . . . n need not have the same resolution nor fixed resolution (e.g.,sensor 23). The interrogation range of sensor 21 is reduced to be withina few frequency increments of the measurement acquired during theprevious sweep. Dynamic measurements can also be produced by comparingvariation in frequencies and amplitudes current sweeps with those of theprior sweeps. For example, if capacitance and inductance are fixed andif the circuit follows a known trajectory (e.g., displacement of alever), the change in position of the sensor 16 is known by comparingthe amplitude variations of successive sweeps. The method requirescalibration to ascertain inductor magnetic response amplitude dependencyto position from antenna 12 (i.e., (A(d))). The calibration correlatesresponse amplitude with distance from the antenna 12. The time betweenmeasurements is ΔT. Hence, displacement rate is derived as displacementrate=[d(A(sweep 1))−d(A(sweep 2))]/ΔT.

Similarly, dynamic strain measurements can be determined by comparingthe frequencies of successive amplitudes. The measurement system canalso be used to identify an amplitude threshold at a set frequency. Thisis indicitive of a certain antenna-inductor separation. If motion isrotary, the rate that the threshold is exceeded (number of times duringa fixed duration) is indicative of rotation rate.

The sweep of individual frequencies is used because it concentrates allenergy used to excite the sensor 16 at a single frequency. FIG. 4depicts a sensor's 16 response amplitude as the excitation frequencyapproaches the sensor's 16 resonant frequency. During each frequencysweep for each sensor 16 range, the current, A_(i), and previous twoamplitudes (A_(i-1) and A_(i-2)) and frequencies are stored. Theamplitudes are compared to identify the amplitude inflection. Thefrequency at which the amplitude inflection occurs is the resonantfrequency. The purpose of the initial sweep is to ascertain all resonantfrequencies and their corresponding amplitudes. Frequencies andamplitude values of successive sweeps can be compared to previous sweepsto ascertain if there is any change to a measured property or if thesensor 16 has moved with respect to the antenna 12. If the physicalstate has changed, the resonant frequency will be different from theprior sweep. If a sensor 16 has moved with respect to the antenna 12,the amplitudes will be different (frequency will remain constant). Themagnitude and sign of the difference can be used to determine how fastthe sensor 16 is moving and whether the sensor 16 is moving toward theantenna 12 or away from the antenna 12.

The interrogation logic can be extended to allow for resistivemeasurements. Once the resonant frequency and its respective amplitudefor a sensor 16 have been identified, the amplitude at a fixed frequencyshift prior to the resonant is then acquired. The resistance isinversely proportional to the difference of the amplitudes. Resistivevariations can be discerned using only two points of the magnetic fieldresponse curve. The bandwidth of the response is proportional to thecircuit resistance. However, to measure bandwidth, it is necessary toidentify the response peak and then measure the response curve on eitherside of the peak to ascertain the 3 dB reductions in amplitude.Identification of the 3 dB reduction would require measuring allamplitudes for each discrete frequency until the reduction amplitudesare identified. Another method to identify characterized resistance isto examine how much the amplitude is reduced from the peak at a fixedfrequency, Δω, separation from the resonant frequency, ω_(r). FIG. 5illustrates response curves for four resistive values. The difference inamplitude between peak response, I₀, and the amplitude at a fixedfrequency away, I(ω*), is inversely proportional to resistance. Thesensor's 16 magnetic field is proportional to its current. The currentat frequency ω* is $\begin{matrix}{{I\left( \omega^{*} \right)} = \frac{ɛ_{0}}{\sqrt{\left( {{\omega^{*}L} - \frac{1}{\omega^{*}C}} \right)^{2} + R^{2}}}} & (1)\end{matrix}$whereω*=ω_(r)−Δω  (2)The amplitude reduction is $\begin{matrix}{{{I\left( \omega_{r} \right)} - {I\left( \omega^{*} \right)}} = {ɛ_{0}\left( {\frac{1}{R} - \frac{1}{\sqrt{S^{*2} + R^{2}}}} \right)}} & (3)\end{matrix}$where $\begin{matrix}{S^{*} = {{\omega^{*}L} - {\frac{1}{\omega^{*}C}.}}} & (4)\end{matrix}$Because $\begin{matrix}{\sqrt{S^{*2} + R^{2}} > {R\quad{and}}} & (5) \\{{\frac{1}{R} > \frac{1}{\sqrt{S^{*2} + R^{2}}}},} & (6)\end{matrix}$the above expression is monotonic with respect to R for fixed S*.Therefore,R=f(I(ω_(r))−I(ω*)).   (7)Equation (7) indicates that resistive measurements can be derived fromthe difference of amplitudes, I(ω_(r))−I(ω*). Once amplitude reductionvariation resistance, Equation (7), has been characterized, this methodrequires only two amplitude measurements to determine resistance, ascompared with the multiple measurements required to determine 3 dBreduction.

The interrogation means comprises hardware for producing a varyingmagnetic field at a prescribed frequency and algorithms for controllingthe magnetic field produced and for analyzing sensor 16 responses. Aschematic of the interrogation system is shown in FIG. 6 and referencedgenerally by the numeral 60. The schematic illustrates the control logicand antenna 12 signals during transmission and reception. Duringtransmission, the microcontroller 605 places antenna 12 intotransmission mode and submits a binary code to frequency synthesizer610. The frequency corresponding to this code is stored in memory 650.The frequency synthesizer 610 converts the code into a square wave, withthe frequency of the wave being dependent on the code. An example of asuitable frequency synthesizer 610 is a DS1085L, made by DallasSemiconductor, which interfaces easily to microcontroller 605 forin-situ programmable frequencies from 4 KHz to 66 Mhz with a controlledresolution of 5 KHz. A high-speed amplifier 615 then amplifies thesquare wave. All frequencies that are higher than the prescribedfrequency are then attenuated using a low pass filter 620. The signal isthen applied to the antenna 12 for a prescribed number of cycles of thewave. The minimum number of cycles should be that required to have thesensor 16 reach it steady-state response amplitude while excited by theantenna 12. The excited steady-state response is dependent upon theantenna 12 frequency, antenna 12 output, sensor 16 resonant frequencyand damping in the sensor 16 due to inherent resistance. There is nomaximum number of cycles. The antenna 12 should remain in thetransmission more long enough to have the sensor 16 reach its steadystate response. The signal to the antenna 12 results in a time varyingmagnetic field. When the cycles are completed or after a set timeduration is completed, the microcontroller 605 switches the antenna 12to receiving mode via a RF receiving/transmission switch 625. During thetransmission, the sensor 16 has current produced in it via Faradayinduction. The sensor's 16 magnetic field decays when the antenna 12 isplaced in the receiving mode. The minimum time duration that the antenna12 should stay in the receiving mode is long enough for the sensor 16 tocomplete at least two cycles of free-decay. The response from the sensor16 is amplified 630 after being received from the antenna 12. A diodepeak detector 635 rectifies the signal (i.e., only the positive value ofthe signal is allowed to pass) and creates a DC value proportional tosignal amplitude (i.e., a capacitor charge is proportional to signalamplitude). An op amp 640 amplifies the DC voltage from the peakdetector 635. The signal from the op amp 640 is then converted into adigital signal, by an A/D converter such as a National SemiconductorADC08831 A/D converter, an eight bit serial analog to digital converterwhich can interface to the microcontroller 605. The microcontroller 605stores the amplitude (digital signal from op amp 640) and thetransmission frequency.

The process described above is iterative for all discrete frequenciesbeginning with the frequency corresponding to the lower bound of thefrequency partition for the sensor 16 with the lowest frequency rangeand continues to the upper bound of the sensor 16 with the highestfrequency range. During the first two iterations of frequency for eachpartition, the amplitudes and frequencies are stored for each sensor 16.

During subsequent iterations, the current amplitude is compared to theprevious two amplitudes to determine if the prior amplitude is aninflection point. Once an inflection amplitude has been detected, theinflection amplitude and frequency are stored, and then the nextpartition is examined. After the last partition is examined, a new sweepis started. Alternatively, during subsequent iterations, the currentamplitude is compared to the stored amplitude. This requires only twostorage locations, frequency and amplitude. If the current amplitude isgreater than the stored amplitude, the current amplitude and frequencyare stored and the previously stored amplitude and frequency arediscarded. No response inflection has been identified and there is ashift to the next transmission frequency in the partition. If thecurrent amplitude is less than the stored amplitude, then the storedamplitude is the response peak amplitude. The transmission frequency isthen shifted to the lower bound frequency of the next partition. If itis the final partition, the transmission frequency is shifted to thefirst partition.

The objective of the aforementioned iterations is to identify theinflection point of each sensor's 16 magnetic field response. Once aninflection amplitude has been detected, the inflection amplitude andfrequency are stored and then the next partition is examined. After thelast partition is examined, a new sweep is started.

A third alternative is to sweep and store all data for the entire rangeif the microcontroller 605 has sufficient memory. Afterwards, peakamplitudes can be ascertained for each sensor 16 partition. The peakamplitudes and their respective frequencies are stored for comparisonsto subsequent sweeps.

The sweep duration must be less than half the Nyquist period of themeasured physical state with the highest frequency. For example, if onesensor is measuring vibrations of less than 30 Hz and other measuredstates have rates of change less than 30 Hz, then the sweeps must bedone at a rate of 60 Hz or greater. All partitions should be examinedduring the first sweep. Subsequent sweeps allow for measurement of timevarying properties. However, subsequent sweeps do not require that allpartitions be examined. The frequency of inclusion of partitions insubsequent sweeps depends upon the desired sampling rate for a givenmeasurement. After the initial sweep, the range of frequencies examinedwithin a given partition can be narrowed to a band of a select number offrequencies on either side of the one identified during the sweep.Narrowing subsequent sweep bands can be used as a means of increasingthe sweep rate. Discrete frequencies need not be evenly spacedthroughout the frequency range (the range includes all sensor 16partitions). However, they should be evenly spaced for each partition.The higher the number of discrete frequencies within a partition, thehigher the sensor 16 resolution.

Each sensor 16 requires a data file that has sensor 16 type, responsevariation, frequency partition and measurement band for each partitionsweep after the resonant is identified on the initial sweep. A tablethat correlates response variation to a physical state value is part ofthe data file. Examples of data files for fluid-level, proximity, androtation sensing are provided below in Tables I, II and III,respectively.

All files are concatenated to form an aggregate file (i.e.,interrogation file=file1, file2, . . . :, file3). Using the examplesgiven above, the aggregate file would be a concatenation of theproximity, fluid, rotation sensor files in the respective order ofincreasing frequency range. The aggregate file is used for regulatingantenna 12 scanning and for converting information acquired during scanto value of physical state.

Additional sensors 16 are added to the system by appending their datafile to the existing aggregate file. Afterwards, a sorting algorithm,such as any of those very well known in the art, is used to sequence thefiles in ascending partition frequency rate. The addition of new sensors16 only requires appending the new sensor's 16 data file to theaggregate data file. No wiring of the sensor 16 to the interrogationsystem is needed nor is there a data acquisition channel dedicated tothe sensor 16. This allows simple implementation of a sensor 16 duringany phase of a system's life or use (e.g., during manufacturing, at timeof part replacement, or during vehicle overhaul). Also important is thatmeasurements of two unrelated physical properties can be derived fromthe same sensor 16 by independently analyzing response amplitude,response frequency or response bandwidth. An example would be strain asone measurement and distance away from a position (e.g. antennalocation) as a second measurement. TABLE I Fluid-level data file LevelFrequency 0 6.837 0.5 6.7915 1 6.7265 1.5 6.6735 2 6.629 2.5 6.5755 36.5155 3.5 6.4605 4 6.414 4.5 6.367 5 6.336 5.5 6.289 6 6.2455 6.5 6.2027 6.1625 7.5 6.1155 8 6.0805 8.5 6.0395 9 5.989Sensor type: fluidResponse variation: FrequencyStart frequency: 7.5 MHzEnd frequency: 5.5 MHzBand: 3

TABLE II Proximity data file Translation Frequency 0.05 4.00E+06 0.0753.63E+06 0.1 3.35E+06 0.125 3.10E+06 0.15 2.89E+06 0.175 2.68E+06 0.22.51E+06 0.225 2.38E+06 0.25 2.27E+06 0.275 2.17E+06 0.3 2.07E+06 0.3251.99E+06 0.35 1.90E+06 0.375 1.83E+06 0.4 1.75E+06Sensor type: ProximityResponse variation: FrequencyStart frequency: 4.5 MHzEnd frequency: 1.5 MHzBand: 3

TABLE III Rotation data file Position Amplitude (Counts) 0 100 90 60 18020 270 60Sensor type: RotationResponse variation: AmplitudeStart frequency: 8.50 MHzEnd frequency: 8.50 MHzParameter Influence

The basic physics of the measurement system will be discussed tohighlight how key parameters influence the magnetic field response ofthe sensor 16 and measurement acquisition. Two simple circuits will beused to aid in the discussion. The first circuit is that of aninterrogating antenna 12 loop of radius a at a distance, r, from thesensor. A harmonic voltage is applied to the loop. The circuit isdesigned to switch from a transmitting antenna to a receiving antenna.During transmission, a harmonic voltage, V, of frequency, ω, is applied.The voltage isV=V_(0 cos ωt.)   (8)The loop has inherent resistance, R_(a), resulting in the loop current,I_(a), being $\begin{matrix}{I_{a} = {\frac{V_{0}}{R_{a}}\quad\cos\quad\omega\quad{t.}}} & (9)\end{matrix}$The current produces a time-varying magnetic field in the circuit. Inthis discussion, the sensor 16 is positioned at a distance r from theantenna 12 plane along the antenna 12 axis. The magnetic field, B, atthe sensor 16 is $\begin{matrix}{B = {\frac{I_{a}\mu\quad a^{2}}{2\left( {a^{2} + r^{2}} \right)^{3/2}}\quad = {\frac{V_{0}}{R_{a}}\frac{\mu\quad a^{2}\cos\quad\omega\quad t}{2\left( {a^{2} + r^{2}} \right)^{3/2}}}}} & (10)\end{matrix}$When r²>>a², the magnetic field is approximately $\begin{matrix}{= {\frac{V_{0}}{R_{a}}\frac{\mu\quad a^{2}\cos\quad{\omega t}}{2r^{3}}}} & (11)\end{matrix}$The permeability, μ, is dependent upon the material that is placed uponthe antenna 12. If nothing is in proximity to the antenna 12 loop, thenthe permeability of free space, μ₀=4π×10⁻⁷ N/ampere² can be used. Thefield is dependent upon the applied voltage, permeability of material incontact with antenna 12, amount of parasitic resistance, antenna 12radius and the distance separating the sensor 16 from the antenna 12.The field strength decays cubically with separation distance.

The second circuit, shown in FIG. 7, is that of the passive sensor 16.To simplify discussion, the sensor 16 is a capacitor c in a seriescircuit. Inductance L and resistance R are inherent to the circuit. Thesecond circuit has a radius r₁. The magnetic flux, Φ_(B), acting uponthe sensor 16 isΦ_(B) =∫B·dS.   (12)Note that B (flux strength and direction) and S (sensor 16 surface areaand normal) are both vector quantities. Maximum flux occurs when theflux and the sensor 16 normal are parallel. Measurements can be acquiredas long as these vectors are not perpendicular. When sensor 16 normaland flux are parallel, the flux is $\begin{matrix}{\Phi_{B} = {\frac{V_{0}}{R_{a}}\pi\quad r_{1}^{2}{\frac{\mu\quad a^{2}\cos\quad\omega\quad t}{2r^{3}}.}}} & (13)\end{matrix}$In accordance with Faraday's law of induction, the induced electromotiveforce, ε, produced in the sensor 16 is equal in magnitude to the ratethat the flux is changing, $\begin{matrix}{ɛ = {- {\frac{\mathbb{d}\Phi_{B}}{\mathbb{d}t}.}}} & (14)\end{matrix}$At the sensor 16, this quantity would be $\begin{matrix}{ɛ = {\frac{V_{0}}{R_{a}}\pi\quad r_{1}^{2}\omega{\frac{\mu\quad a^{2}\quad\sin\quad\omega\quad t}{2r^{3}}.}}} & (15)\end{matrix}$When the antenna's 12 magnetic field is harmonic, the resultingelectromotive force produced in the sensor 16 is dependent upon flux,the area of sensor's 16 inductor and is proportional to the frequency ofthe flux. The constituent components of the sensor 16 are in series. Thedynamics of current in the sensor 16 is $\begin{matrix}{{{{LI}^{\prime} + {RI} + {\frac{1}{C}{\int{I\quad{\mathbb{d}t}}}}} = {ɛ_{0}\sin\quad\omega\quad t}}\quad} & (16)\end{matrix}$with $\begin{matrix}{ɛ_{0} = {\frac{V_{0}}{R_{a}}\pi\quad r_{1}^{2}\omega\frac{\mu\quad a^{2}}{2r^{3}}}} & (17)\end{matrix}$and L, R, C and I, are the sensor's 16 inherent inductance, inherentresistance, capacitance and current. Equation (16) is differentiated toeliminate the integral, resulting in $\begin{matrix}{{{LI}^{''} + {RI}^{\prime} + {\frac{1}{C}I}} = {\omega\quad ɛ_{0}\cos\quad\omega\quad{t.}}} & (18)\end{matrix}$The solution of Equation (18) is $\begin{matrix}{{{I_{TX}(t)} = {\frac{ɛ_{0}}{\left( {S^{2} + R^{2}} \right)}\left\lbrack {\frac{{\left( {{\lambda_{2}S} - {R\quad\omega}} \right){\mathbb{e}}^{\lambda_{1}t}} + {\left( {{R\quad\omega} - {\lambda_{1}S}} \right){\mathbb{e}}^{\lambda_{2}t}}}{\left( {\lambda_{1} - \lambda_{2}} \right)} + {S\quad\cos\quad\omega\quad t} - {R\quad\sin\quad\omega\quad t}} \right\rbrack}}{with}} & (19) \\{S = \left( {{\omega\quad L} - \frac{1}{\omega\quad C}} \right)} & (20) \\{\lambda_{1} = {{- \frac{R}{2L}} + {\frac{1}{2L}\sqrt{R^{2} - \frac{4L}{C}}}}} & (21) \\{\lambda_{2} = {{- \frac{R}{2L}} - {\frac{1}{2L}{\sqrt{R^{2} - \frac{4L}{C}}.}}}} & (22)\end{matrix}$The subscript, _(TX), denotes that the antenna 12 is transmitting. Theterm, S, is reactance.

The sensor 16 current when the antenna 12 is transmitting is given byEquation (19). The steady state response of the sensor's 16 currentwhile the antenna 12 is transmitting is $\begin{matrix}{{{I_{p}(t)} = {I_{0}{\sin\left( {{\omega\quad t} \pm \theta} \right)}}}{where}} & (23) \\{{I_{0} = \frac{ɛ_{0}}{\sqrt{S^{2} + R^{2}}}}{and}} & (24) \\{{\tan\quad\theta} = {\pm {\frac{S}{R}.}}} & (25)\end{matrix}$The term {square root}{square root over (S²+R²)} is impedance.

Equation (24) has the influence of sensor's 16 resistance, reactance andelectromotive force level on the steady current amplitude, I₀, when theantenna 12 is transmitting. It can be concluded by examination ofEquation (24), that the amplitude is maximized by minimizing resistanceand reactance. Resistance is minimized by increasing electricalefficiency of constituent components. Reactance is zero when the antenna12 broadcast frequency is that of the undamped resonance of theinductive-capacitive circuit, which is $\begin{matrix}{\omega = {\frac{1}{\sqrt{LC}}.}} & (26)\end{matrix}$The time to reach steady state is dominated by the larger of the tworoots, λ₁. As can be seen from the root, the decay rate is proportionalto resistance and inversely proportional to inductance. After a finiteamount of time, Δt, the interrogation antenna 12 is switched to thereceiving mode, thus removing the electromotive force from the sensor16. The sensor 16 current response is now $\begin{matrix}{{{LI}^{''} + {RI}^{\prime} + {\frac{1}{C}I}} = 0.} & (27)\end{matrix}$The response is overdamped if ${R^{2} > \frac{4L}{C}},$critically damped if ${R^{2} = \frac{4L}{C}},$or underdamped if $R^{2} < {\frac{4L}{C}.}$The overdamped response could occur if a resistive type measurement isadded to the circuit and inductance and capacitance are kept constant.If an operational objective is to have considerable separation distancebetween the sensor 16 and the antenna 12, then the sensor 16 should onlybe composed of capacitive and inductive elements. If possible, thesensor 16 should be designed to reduce inherent resistance. The solutionfor the underdamped case is $\begin{matrix}{\begin{matrix}{{I_{RX}(t)} = {{\mathbb{e}}^{\frac{- R}{2L}{({t - {\Delta\quad t}})}}\left\lbrack {{A\quad\cos\left( {\sqrt{\frac{1}{LC} - \frac{R^{2}}{4L^{2}}}\left( {t - {\Delta\quad t}} \right)} \right)} +} \right.}} \\\left. {B\quad\sin\left( {\sqrt{\frac{1}{LC} - \frac{R^{2}}{4L^{2}}}\left( {t - {\Delta\quad t}} \right)} \right)} \right\rbrack\end{matrix} =} & (28) \\{{I_{RX}\left( {\Delta\quad t} \right)} = {{I_{TX}\left( {\Delta\quad t} \right)} = {A.}}} & (29)\end{matrix}$The subscript, _(RX), denotes that the antenna 12 is receiving.

The decay envelop depends on −R/2L. The current value in the sensor 16,I_(TX)(Δt), when the antenna 12 is switched to receiving mode andcurrent derivative value, I′_(TX)(Δt), are the initial conditions usedto determine coefficients A and B. In a manner similar to the antenna12, the magnetic field produced by the sensor 16 is now $\begin{matrix}{B_{RX} = {{\frac{{I_{RX}(t)}\mu\quad r_{1}^{2}}{2r^{3}}\quad{for}\quad r^{2}} ⪢ {r_{1}^{2}.}}} & (30)\end{matrix}$As can be seen by Equation (30), the magnetic field is dependent uponthe sensor's 16 current, which is dependent upon the electromotiveforce, reactance and resistance.

During subsequent transmission intervals, the final conditions from theprior mode (e.g., transmission or reception) are the initial conditionsfor the current mode. Hence, each transmission and reception intervalhas a closed form solution for current response. Table IV summarizes theinfluences of various parameters on the sensor's 16 magnetic fieldresponse. TABLE IV Influence of parameters on sensor response Effect onSensor's Magnetic Field Response Parameter when Parameter is IncreasedAntenna voltage Amplitude increases Antenna inherent Amplitudedecreases; increasing width of resistance antenna trace reducesresistance Permeability of Amplitude increases material in contact toantenna Antenna diameter Amplitude increases Antenna-sensor Amplitudedecreases cubically separation Sensor orientation Amplitude maximizedwhen sensor normal and with respect magnetic flux are parallel and zerodegrees with flux from antenna perpendicular Sensor inductance Amplitudeincreases area Frequency of antenna Amplitude increases magnetic fieldReactance Amplitude decreases; amplitude maximized when antennafrequency tuned to sensor circuit frequency Sensor inherent Amplitudedecreases, bandwidth increases; resistance or applied increasing widthof inductor trace reduces resistance resistance Ratio of sensor Sensorresponse decay rate increases as ratio is resistance to sensordecreased. inductance

The distance at which the magnetic inductor response can be received isproportional to the strength of the magnetic field created in theinductor. The magnetic field strength is dependent upon the current inthe sensor 16. Therefore, interrogation distance is also dependent uponthe energy efficiency of the sensor 16. The higher the energyefficiency, the more current is created for the same level of power usedby the interrogating antenna 12. The quality factor, Q, isrepresentative of this efficiency. Q is the ratio of reactance to DCresistance. A stronger magnetic field is created with higher Q.

A magnetic field response sensor 16 is metamorphic if a physicalproperty for which it measures, or if its environment, results in apermanent non-reversible change in one or more of its constituentcomponents. The change results in a new reference (i.e., baseline)magnetic field response, thus giving it the ability to make othermeasurements. Examples of metamorphic changes include chemical reactionor phase transition and strain experienced during yield or cracking.Dielectric or permeability changes resulting from a phase transition,such as resin curing or chemical reactions, produce irreversible changesto a sensor 16. If interdigital electrodes are used for capacitance, thesensor is capable of measuring strain, displacement, or another physicalproperty after the dielectric changes. During the dielectric change, asensor 16 can be used to track the change (e.g., rate of curing oramount of chemical reaction). A new response baseline results from thecompleted dielectric change. A sensor 16 (e.g., a spiral inductor andinterdigital capacitor) for measuring strain can be affixed to a surfacevia a direct metal deposition method. Direct deposition of a metallicthin film does not add any increased structural integrity to thesurface. If a crack forms on the surface along the capacitor, causingsome but not all of the capacitors to be severed, the sensor is stillcapable of determining other measurements (e.g., displacement). After acrack, strain can still be discerned, but referenced to a differentbaseline frequency. Other examples include a permanent structural yieldto one of the components as a result of excessive strain. Sensormetamorphosis allows measurement of a physical property that undergoesan irreversible change to transform the sensor 16 into a means ofmeasuring other physical states.

The acquisition system, the sensor 16 and the immediate environment ofthe sensor 16 form a triad. Unlike traditional sensors, a unique featureof magnetic field response sensors is that, when used with theinterrogation system described herein, they can easily be transformedfrom the means of measuring one physical state to measuring that ofanother physical state. The magnetic field response of the sensor 16 isthe means of acquiring the measurement from the sensor 16. The field canbe varied by changes to multiple physical states influencing the sensor.Each constituent of the sensor 16 can be used for measurement.Capacitive variations result in sensor 16 response frequency variations.Inductive variations can result from position variations to conductivesurfaces. The position variations change both frequency and amplitude ofthe sensor 16 response. When a sensor's 16 constituent values remainfixed, changes to the antenna/sensor separation produce an inversevariation of response amplitude. The aforementioned variations result inchanges to response frequency, amplitude or both. Because they areindependent, a single sensing element can be used to measure more thatone independent physical property. A valid measurement is achieved byfixing all but one physical state. The variable state is the measuredstate. Table V summarizes changes to a sensor's physical orenvironmental attributes and the subsequent response change. Metaphoricsensors are also multi-functional, except that the irreversible propertyfor which they measure can only be measured once. TABLE V Changes tosensor magnetic field response due to parameter variation SensorVariation Attribute(s) of Magnetic Field Response Inductor positionrelative to conductive surface Amplitude decreases (dA), frequencydecreasing increases (dω) Inductor surface area overlap of conductiveAmplitude decreases (dA), frequency surface increasing increases (dω)Capacitor plates or electrodes separation Frequency decreases (dω)decreasing Capacitor plates area overlap increasing Frequency decreases(dω) Dielectric immersion of electrodes or Frequency decreases (dω)electroplates increases Increased electrodes (e.g., electrical contactof Frequency decreased (dω) two interdigital capacitors) Increasedinductance (e.g., electrical contact of Frequency increased (dω) twoinductors) Dielectric phase transitions Frequency change (increase ordecrease depends upon electrical properties of each phase) Dielectricchange due to chemical reaction Frequency change (increase or decreasedepends upon electrical properties of each phase) Dielectric change dueto environmental Frequency increased (dω) exposure Inductor distancefrom antenna increases Response amplitude decreases (dA) Resistance incircuit increases Amplitude decreases (dA) and bandwidth increases (df)

The states need not have any relation to each other. An example ofmulti-functional sensing would be a sensor that uses interdigitalelectrodes for a capacitor embedded in a tire prior to curing rubber.The sensor has an initial response baseline. During curing, thefrequency changes due to the material phase transition. Once cured, anew response baseline is established. Deformations or pressurevariations to the tire change the spacing between electrodes and thusresult in perturbations to the baseline frequency. This measurement istaken prior to motion of the vehicle, thus updating the responsebaseline for rotation measurements. If the antenna used to interrogatethe sensor maintains a constant position and orientation, the rotationof the tire results in the amplitude of the response varying between twolevels. The rate that the amplitude varies is the rate that the tirerotates. This example demonstrates that a single sensor can be used formeasuring three independent properties: 1) tire curing 2) tirepressure/deformation and 3) tire rotation. If the tires are steel beltedand the sensor is placed on the inside wall of the tire at a fixedseparation from the steel belts, any change in inductor positionrelative to the steel belts could be indicative of tire ply separation.Under these conditions, a fourth measurement, bond separation, isachieved.

The manner in which a sensor 16 is interrogated and the responsebaselines updated allows for metamorphic and/or multi-functional use ofmagnetic field response sensors 16. Another example of multiplemeasurements being derived from a single sensor 16 is that of a movinglinkage or door. Consider a lamina-type sensor 16 that is attached to adoor for which knowledge of contact to another surface and its motion isrequired. The knowledge of contact is achieved by electrically shortingthe sensor 16 with the contact surface. Knowledge of motion is achievedby examining the amplitude of the response. Further, the capacitiveelement could be used to measure other properties, such a strain ormoisture.

Metamorphic sensors are interrogated in the same manner describedearlier. When the permanent change of the sensor 16 is complete, a datafile for measurements of the transformed sensor need to be concatenatedto the aggregate file used for regulating the sensor 16.

Measurements from multi-functional sensors can be analyzed in twomanners. In one embodiment, the sensor 16 measures one physical stateand then is returned to its baseline frequency to measure the otherstate. When sensors are returned to their baseline, a separate data filefor each type of measurement is required. The frequency partitions foreach use will have some degree of overlap. Ideally, the sensor 16returns to its baseline. If the sensor 16 cannot be returned to itsbaseline prior to measuring a second state, then a correlation tablesuch as Table VI needs to be developed. The table allows combinations ofamplitude and frequency to be correlated to combinations of physicalvalues for State X and State Y. As can be seen from Table VI, if theamplitude and frequency, A₃, ω₃, are the sensor baseline, the thirdcolumn of combinations in Table VI would be the correlation data forState X. Similarly, the third row would be the correlation data forState Y. TABLE VI Multi-functional sensor correlation table Variation ofPhysical State Y State Y₁ State Y₂ State Y₃ State Y₄ State Y₅ Variationof State X₁ A₁, ω₁ A₂, ω₁ A₃, ω₁ A₄, ω₁ A₅, ω₁ Physical State State X₂A₁, ω₂ A₂, ω₂ A₃, ω₂ A₄, ω₂ A₅, ω₂ X State X₃ A₁, ω₃ A₂, ω₃ A₃, ω₃ A₄,ω₃ A₅, ω₃ State X₄ A₁, ω₄ A₂, ω₄ A₃, ω₄ A₄, ω₄ A₅, ω₄ State X₅ A₁, ω₅A₂, ω₅ A₃, ω₅ A₄, ω₅ A₅, ω₅

In cases where rotation rate is one measurement and a thresholdamplitude is used for determining rotation (i.e., the numbers of timesthe threshold is exceeded per minute), only the correlation informationfrom the other measurement is required. If the other physical state'srate of change is far less than the rotation rate, the amplitudethreshold could be a certain percent of the physical state's lastmeasured amplitude.

Antenna Design

Parametric measurements were performed to ascertain the influence ofgeometric properties on interrogation antenna effectiveness, i.e.,energy efficiency. To facilitate non-obtrusive use of the measurementsystem, the antennae were developed as either thin-film deposited on adielectric membrane or thin foil which can be placed on any existingnon-conductive surface. To ascertain the effect that geometry would haveon the electrical properties, two features were considered: antennawidth and antenna diameter. FIG. 8 is representative of a thin copperfoil antenna 12 adhered to a Plexiglas plate. The antenna 12 trace widthwas initially 2.0 in. In a first study, the antenna's outer diameterremained a constant 18.0 inches. The inner diameter was reduced andmeasurements were taken when the antenna trace was 2.0, 1.5, 1.0, 0.5,and 0.25 in. The inductance, DC resistance and Q were measured for eachwidth. A current of 1 KHz was used for the inductance measurements andthe Q measurements.

For the second antenna used for the parametric measurements, six 0.5 intraces of copper foil were adherred to a Plexiglas plate. The outerdiameters were 6, 8, 10, 12, 14 and 16 inches. Coaxial cable wasindividually electrically connected to each trace. The inductance, DCresistance and Q were measured for each width. Resistance measurementsare shown in FIG. 9 for both parametric changes to antenna width anddiameter. As seen in FIG. 9 a, DC resistance decreased with increasedtrace width. Resistance increased significantly as the width wasreduced. The resistance changed from 0.052 Ω to 0.118 Ω as the width waschanged from 0.5 in to 0.25 in. The resistances of the wider traces weresubstantially less. For the traces wider than 1.0 in, the resistancedecreased to a lesser extent. As seen in FIG. 9 b, the resistanceincreased approximately linearly with diameter. The measurement resultsindicate that to develop low resistance antennae, a wide trace wouldresult in less applied power loss due to lower resistance.

Inductance measurements are presented in FIGS. 10 a and 10 b. Thesemeasurements have similar trends as the resistance measurements.Inductance increases are more pronounced for narrower traces. Inductancealso increases approximately linearly with increasing diameter. Valuesof Q are presented in FIGS. 11 a and 11 b. An antenna's electricalefficiency is dependent upon its Q (i.e., higher Q results in higherefficiency). The trace width has a significant effect on Q. The increaseof Q with increasing trace width is approximately linear. As the widthwas changed from 0.25 in to 2.0 in, Q changed by greater than a factorof 4, as shown in FIG. 11 a. Changing the outer diameter from 6 in to 16in resulted in Q changing by less than 0.02, as shown in FIG. 11 b.

Inductor Design

The effect that design features such as perimeter size and trace widthhad on inductance, DC resistance and Q was examined. The inductor servesto relay the measurement. The distance at which the magnetic inductorresponse can be received is proportional to the strength of the magneticfield created in the inductor. The magnetic field strength is dependentupon the current in the sensor 16. For the same applied energy, a lowerresistance results in a higher current. Hence, to increase the range ofthe sensor 16, the sensor elements should have as low of a resistance aspossible.

FIG. 13 illustrates the effect of trace width on DC resistance. Three3-inch square spiral inductors, as shown in FIG. 12, having widths 120of 0.02, 0.25 and 0.50 inches were used. The resistance of the 0.02 intrace was 9.4 Ω. Resistances for the 0.25 and 0.5 in traces were 0.055 Ωand 0.023 Ω, respectively. The significantly lower resistance of thewider traces demonstrates that the trace width is an effective designparameter. The effects of trace width and inductor perimeter size on Qare shown in FIG. 14. The values of Q for 3 in (0.02, 0.25, and 0.50widths) and 5 in square spirals (0.25 in and 0.20 in widths) are shown.The value of Q increases approximately linearly with increasing tracewidth. The larger size square spiral results in a higher Q for the sametrace width. Comparison of the 5 in square having the 0.25 in trace withthe 3 in square spiral having the 0.50 inch trace shows that increasingwidth can be used as a method of producing a higher Q.

To quantify effective range for measurement acquisition, the inductorswere coupled to capacitors. Two measurement configurations wereinvestigated. In the first configuration, a switching antenna (12 inouter diameter loop using 12 gauge copper wire) was used with atransmission power level of 0.1 W. An inductor with a 5 in ×5 in squarespiral with a 0.75 in trace, coupled to a 504-pF capacitor, achieved a−60 dB response at a 25 in distance from the antenna. The inductor witha 3 in×3 in square spiral with a 0.25 in trace, coupled with a 826-pFcapacitor, achieved a −6 dB response at a 22 in distance from theantenna. In a second measurement configuration, a transmission antenna(18 in outer diameter and 0.5 in trace) and separate receiving antenna(24 in outer diameter using 12 gauge copper wire) were used. They werepositioned 11 ft apart. The antennae were operated such that thereceiving antenna was off when the transmission antenna was powered onto excite the sensors 16. The transmission antenna used 1.5 W of power.When the transmission antenna was switched off, the receiving antennawas powered on, allowing it to receive the sensor's 16 response. In thisconfiguration, the sensing elements could be interrogated anywhere in avolume approximated by a cylinder whose longitudinal axis ran betweenthe antennae centers and with a diameter of approximately 4 ft. Thelength of the cylinder was the separation distance between the antennae.When the antennae were separated by 9 ft, the same sensing elementscould be interrogated using 1.0 W of power. Using a single antennaelectrically switched from a transmitting to receiving antenna, aninterrogation distance of 2 ft was achieved using 0.1 W of power appliedto the antenna.

It is necessary in some applications to have the sensor's 16 capacitoraffixed to or embedded in a conductive surface. Proximity to conductivesurfaces alters the inductance of the sensors. As the sensor gets closerto a conductive surface, the magnetic field energy of the sensor isreduced due to eddy currents being induced in the conductive surface.The inductor cannot be affixed to or embedded in a conductive surface.It is necessary to have a means of fixed separation (at least 0.375 in).The minimum distance for separation is determined by the sensor 16response. The inductor should be separated from the conductive surfaceso that the response amplitude exceeds the noise level by a recommended10 dB. FIGS. 15-16 illustrate embodiments for maintaining constantinductance levels. In FIG. 15, a nonconductive dielectric spacer 152 isused to maintain a fixed separation between inductor 155 and conductivesurface 151. Nonconductive film 153 and capacitor 154 are alsoillustrated. Although the inductance is less than what it would be if itwere not in proximity to conductive surface 151, the inductance isfixed. As long as the inductance is a fixed, all variations of themagnetic field response are due to capacitance changes. FIG. 16illustrates a sensor in which the inductor 155 is positioned at a fixedangle away from the conductive surface 151. A lightweight stiffener 156is used to maintain the angle.

Numerous variations of inductor mounting can be utilized, such ashousing that provide separation from the conductive surface as well asprotection from impact damage. Systems that have limited space butundergo deployment can have inductors that deploy during deployment ofthe system and maintain fixed position after deployment is complete,including both rotational and telescopic deployable inductors. Ifcapacitance is maintained fixed in value, changes in inductanceresulting from variation of the separation between inductor andconductive surface can be used to measure proximity to that surface.This variation depends on the surface skin depth.

Table VII illustrates various ways in which variations to thecapacitor's geometric properties can be used for sensing. Plateseparation, plate apparent overlap and the orientation of the platesrelative to each other can be extended to provide a variety ofmeasurements predicated upon the plates' relative change in orientationor position with respect to each other. When interdigital electrodes areused as the capacitor of the sensor, spacing between the electrodes canbe used for sensing. Table VIII illustrates the measurement applicationsresulting from capacitance variation. Table IX illustrates themeasurement applications resulting from dielectric variation. Table Xillustrates various ways in which the variations in either the sensor'sinductance or variation in the sensor field response amplitude can beused for measurements.

Piezoelectric material can be used for the sensor's capacitivecomponent. Piezoelectric materials (e.g., piezo-ceramics such as leadzirconate-titanate (PZT), or piezo-polymers such aspolyvinlydinofloruride (PVDF)) have electrical properties similar tocapacitors. These materials develop electric polarization when force isapplied along certain directions. The magnitude of polarization isproportional to the force (within certain limits). The capacitancevaries as the polarization varies, which suffices for measuringresulting strain from material deformation. Deformation can be due toeither mechanical or thermal loading (pyroelectric effect). Thesematerials can be used in lieu of capacitors for strain and temperaturemeasurements.

The L-C circuit can be directly deposited onto a surface as a thin filmusing photo-lithography. In one embodiment, if the surface isnonconductive, the inductor and interdigital electrodes can be depositedfirst. A layer of dielectric material such as Silicon Nitride (Si3N4)with four electrical vias is deposited next. A via is placed at eachterminus of the inductor and capacitor. A layer having two electricalconduits (trace of conductive material) is then deposited. The twoconduits are position such that they complete the inductor-capacitorelectrical connection. Silicon Nitride also can be used as thin filmcoating for environmental protection of sensor. These dielectric layerscan be deposited by APCVD (Atmospheric Chemical Vapor TABLE VIIMeasurement applications for capacitor geometric variation CapacitiveGeometric Variation Measurement Application Plate separation Proximitysensing - Each plate can be attached to a separate surface. Pressure -If plates are elastic, surface deformation due to external pressurealters separation distance between plates. Strain If a dielectric ofknown elastic modulus is affixed to and between the rigid plates (e.g.,embedded), compression and tension can be measured. If each plate isattached separately and perpendicular to a surface of known elasticmodulus, surface compression and tension can be measured. Apparent area(i.e., Position displacement when no dielectric is used and one plate isfree plate overlap) to move relative to other plate; all plate motionmust be parallel. Shear force When an elastic dielectric of known shearmodulus is affixed to and between both plates (e.g., embedded), shearforce is inversely proportional to plate overlap as plates translatewith respect to each other. As the overlapped area of the plates change,the electric field changes. The electric field exists only within thearea for which the plates overlap. All plate motion must be parallel. Ifeach plate is attached separately and parallel to two surfaces, anyshear force between surfaces is inversely proportional to surfaceoverlap. Torsion - Can also be measured in a somewhat similar manner asshear. In torsion measurements, one plate is rotated about its normalrelative to other plate. Relative plate Angular orientation -Orientation of one plate relative to the other. orientation Onlyapplicable for a single axis of rotation.

TABLE VIII Measurement applications resulting from capacitive variationCapacitive Variation Measurement Application Separation between Strain -In plane strain changes the distance between neighboring neighboringelectrodes electrodes resulting in a change to electric field and thus acapacitance change. Pressure (Vacuum) sensor - Interdigital electrodesare deposited on an elastic dielectric membrane. The membrane is securedto a frame. The frame serves to separate the inductor from theconductive surface and serves as a portion of the cavity that maintainsthe pressure(vacuum). The other surfaces forming the cavity to which thepressure is maintained are the conductive surface and the membrane. Whenthe sensor is exposed to pressure (vacuum) the membrane will deformtoward (away) from the conductive surface thus changing the capacitance.Number of electrodes Numeric encoding - The relative number ofinterdigital electrodes can be used for numeric coding. The use ofinterdigital electrodes allows flexibility in selecting base (e.g.,binary, octal, decimal, hexadecimal, etc) for numeric coding. Forexample (using base 10), a single number can be developed as a singleinductor in parallel with ten electrode pairs. This circuit is theequivalent of a single digit. When more than one digit is needed,similar circuits can be used but different inductance levels are used todistinguish the digits. Ten electrode pairs give the circuitry theability to be resolved as a base 10 numeral. All capacitors have thesame capacitance. A numeral is determined by the number of activeelectrode pairs (i.e., in the non- opened portion of the circuit). Thenumber that the circuit represents is the number of active electrodepairs subtracted from ten. Different inductance values are used whenother digits are needed.

TABLE IX Measurement applications resulting from dielectric variationDielectric Variation Measurement Application Dielectric immersionDielectric level (e.g., fluid level or solid particle level). The sensorresonant changes inversely to dielectric immersion. When interdigitalelectrodes are used, resonant changes discretely with immersion.Dielectric phase Material phase transition (e.g., solid to liquid)changes Reversible Moisture, chemical exposure resulting inenvironmental nonstoichemtric changes to dielectric exposureStoichemetric Examples are hydrogen exposed to a palladium (chemical)changes to dielectric (a means of developing an hydrogen detector) ordielectrics a silicon dielectric exposed to oxygen (a means ofdeveloping an oxygen detector). Each example alters the dielectricproperties.

Deposition)/or LPCVD (Low Pressure Chemical Vapor Deposition)/or PECVD(Plasma Enhanced Chemical Vapor Deposition)/Sputtering/Sol-Gel/Electronbeam lithography/Thermal evaporation/or Microwave methods.Characteristics of silicon nitride can be varied by different gas dopinglike leaking small quantity of oxygen during deposition, or byimplanting nitrogen ions in already deposited silicon nitride. Withvarying doping level and species, refractive index and othercharacteristics of thin film can be varied hence usage for differentapplications. After deposition of silicon nitride film, these films canbe thermally rapid annealed. Furthermore, the capacitor can be directlydeposited upon conductive directly after a dielectrical material as beendeposited upon the surface. The inductor must be spaced or position suchthat its inductance reminds constant. TABLE X Measurement applicationsresulting from inductive variation Inductive Variation MeasurementApplication Inductor proximity to Inductance changes as distance to aconductive surface varies due to conductive surface eddy currents beingproduced in the conductive surface. As an inductor is moved closer tosurface, amplitude decreases and frequency increases. Applications are:Proximity measurement - inductance changes as inductor get closer to aconductive surface. Wear measurement - Inductor is placed on the uppersurface of a material whose thickness is lessen with wear and the lowersurface is in contact with a conductive material. As material wears,inductance decreases due to increased proximity to conductive material.Bond separation - Placement of a conductive surface on one side of asurface bond and a lamina-type L-C element on the other side of a bondsuch that the conductive surface and the L-C element are in proximity toeach other. If the bond is broken, the inductance will change. Anexample would be that for steel-belted tires. If a L-C element is placeon the inside wall of the tire, any separation of the steel belts fromthe rubber would result in an inductance change. Pressure (Vacuum)sensor - Spiral inductor is deposited on an elastic dielectric membrane.The membrane is secured to a frame. The frame serves to separate theinductor from the conductive surface and serves as a portion of thecavity that maintains the pressure(vacuum). The other surfaces formingthe cavity to which the pressure is maintained are the conductivesurface and the membrane. When the sensor is exposed to pressure(vacuum) the membrane will deform toward (away) from the conductivesurface thus changing the inductance. Load sensing - If a material ofknown elastic modulus if affixed to the conducting surface and theinductor surface, axial compression or tension can be measured.Identifying conductive materials - Skin depths for seawater and graphiteare 200 and 1.59 mm at 1 MHz. Aluminum, chromium, copper, gold andsilver have skin depths of 0.085, 0.081, 0.066, 0.075 and 0.064 mm,respectively. The inductance of the sensor is proportional to itsinduced magnetic field. The field (and inductance) decreases as theinductor distance to the conductive surface decreases. As inductancedecreases, the sensor resonant frequency increases. The responseamplitude also decreases as the inductor gets closer to the conductivesurface due to more energy being lost to the conductive material. Theamplitude decay with respect to increased frequency is proportional toskin depths. Therefore, the slope, dA/dω, can be used as a means ofdiscerning water, graphite and metals from each other. Variation ininductor Inductance changes with proximity to a conductive surface thatresults surface area overlap of in L-C amplitude and frequencyvariation. When distance separating conductive material inductor andconductive surface is fixed, the amount of inductance is proportional tothe area overlap of inductor and conductive surface. In a manner similarto capacitive plate overlap variation, one surface has a conductivematerial and the other has the inductor. Applications are: Position anddisplacement measurements. Shear load measurement Torsion loadmeasurements Relative plate orientation Inductor distance from Whencapacitance and inductance are fixed, amplitude of response is receivingand dependent upon distance from receiving antenna and transmittingtransmitting antenna. Both antennae must have fixed position andorientation. antenna(e) Response frequency will not vary but responseamplitude will vary as inductor's position relative antenna(e) changes.Applications are displacement and displacement rate measurements such astire rotation, motion of a linkage, etc. Numeric encoding The relativenumber of inductors in series can be used for numeric coding in a mannersimilar to that for the interdigital capacitor.Specific sensor embodiments

EXAMPLE 1 Sensing Element for Closed Cavities Having Low RFTransmissivity

Examples of closed cavities for which measurements are desired within acavity include metal fuel tanks and landing gear struts. Metalenclosures have low transmissivity for the RF energy. The magnetic fieldproduced from an electrically active inductor is eliminated when placedin very close proximity to an electrically conductive material. Thismeans that antennae or inductors can not be placed on the surface of anelectrically conductive material or embedded in electrically conductivecomposite materials (e.g., graphite fibers). Additionally, to use aconductive material to support an antenna made from metal foil or metaldeposited on a thin film, the antenna must be separated, such as by useof a spacer. The thickness of the spacer is dependent on the amount offield strength that the antenna can lose without losing its ability toacquire its measurement. The same is true for the inductor used in thesensor. If the sensor is placed on a conductive surface, the capacitorcan be placed in contact with the surface (a dielectric layer mustseparate the capacitor and the surface), but the inductor must beseparated from the surface via a spacer. Similarly, the capacitor can beembedded within conductive composite layers but the inductor must beplaced on the outside and separated.

When the cavity containing the sensor 16 is made of a conductivematerial and the antenna 12 is external to the cavity, the inductor mustalso be external to the cavity to allow the sensor 16 to be exposed tothe antenna's 12 varying magnetic field. The inductor must be maintainedin a fixed position relative to and separated from the conductivesurface.

A representative embodiment is shown in FIG. 17. The capacitive element170 of the sensor 16 is situated in a closed cavity 171 and theinductive element 172 of the sensor 16 is placed outside of the closedcavity. This allows the inductive portion 172 of the sensor to radiatein essentially open space and transmit the information gathered by theenclosed capacitive element 170. A broadband antenna broadcastselectromagnetic energy within the frequency range of the sensor andreceives the emissions of the sensor 16, which signals are processed toidentify phenomena associated with each sensor 16.

Referring to FIG. 18, a narrow throat portion 180 of the sensor 16connects the inductor 172 to the capacitor 170. The throat 180 is ofsufficient length to allow the capacitor 170 to be appropriately placedwithin the cavity 171. The inductor 172 is placed outside the cavity171, and separated from the cavity wall 174 via nonconductive spacer176. The throat 180 is fed through the orifice 173 in the cavity wall174 that is used to fill the cavity 171 (e.g., fuel tank opening) andconnects the inductor 172 and capacitor 170 via electrical leads 175 toform a parallel circuit. Another embodiment is to have the inductor 172and capacitor 170 fabricated as separate units. In this embodiment, theinductor 172 is mounted external to the cavity 170 and the capacitor 170is mounted internal to the cavity 171. Electrical leads 175 are fedthrough the orifice 170 that is used to fill the cavity 171 and connectthe inductor 172 and capacitor 170 to form a parallel circuit.

Referring to FIG. 19, when a cavity 171 containing multiple sensors 16is made of a conductive material, an antenna 12 can be placed internalto cavity 171. An internal antenna 12 allows all components of thesensors 16 to reside inside the cavity 171. The antenna 12 must beseparated from the conductive surface 174. The inductors must bemaintained in a fixed position relative to and separated from theconductive surface 174. Antenna leads 190 feed through orifice 173.

EXAMPLE 2 Sensing Element for Material Phase Transition and StrainMeasurement

FIG. 20 a illustrates a sensor embodiment used to measure material phasetransition. The inductor 200 is formed as a square spiral trace ofcopper. Interdigital electrodes are used for the capacitor 202. Theinductor 200 and the capacitor 202 are deposited on a thin dielectricfilm. A single antenna 12 is used to power the sensor 16 and to receiveits response. The resonant frequency of the sensor 16 is 5.6 MHz. As anexperimental example, the sensor 16 was placed in the bottom of aplastic container. Liquid resin was poured into the container and becamea dielectric of the capacitor 202, resulting in the sensor 16 resonantfrequency changing to 4.8 MHz. As the resin cured, its dielectricconstant changed, resulting in a changed capacitive value of the sensor16. FIG. 20 b is an embodiment that distinguishes parts during curing.The circuit can be programmed easily to have a response range for onepart (to be cured) different from another. FIG. 21 shows a time historyof the magnetic field response resonant frequency during resin curing.As seen in FIG. 21, the response frequency had no further change after100 minutes, when the curing was complete. This embodiment can also beused for strain measurements. When the sensor 16 is affixed to asurface, the separation, d, between electrodes will change when thesurface is strained. As the separation changes, the capacitance, andthus the resonant frequency, of the circuit changes.

EXAMPLE 3 Sensing Element for Wear and Thermal Measurements

Applications for sensors 16 which measure wear or temperature includelanding gear or automotive brakes. The sensors 16 can incorporate eitherthe individual functions of wear and temperature measurement or bothcombined. A first embodiment utilizes one or more interdigitalelectrodes 220 positioned such that the electrodes 220 are parallel tothe surface of wear, as illustrated in FIG. 22. The metal used for theelectrodes 220 is a metal that can wear away more easily than thesurface for which wear is to be measured. The device is positioned whilethe volume (of which one surface is to have its wear measured) isliquid. The liquid is cured to a solid, thereby embedding the capacitiveelement. Furthermore, the curing of the material can be monitored. Afterthe material is cured, the sensor can be used for wear measurements. Asthe surface wears away, the primary electrical buses wear away. As wearincreases, electrodes 220 are severed from the bus, thereby altering thecapacitance of the device. FIG. 23 illustrates an embodiment of aninterdigital device with one of the electrodes 230 having a temperaturesensitive dielectric or a dielectric which has a phase transition (i.e.,solid to liquid) when exposed to excessive temperature. The phasetransition dielectric solidifies when the temperature is reduced belowcritical. Hence, it has the function of wear measurement and excessivetemperature indications. When a temperature sensitive dielectric isused, the capacitance changes proportionally with temperature. When aphase transition dielectric is used, the capacitance changes moredramatically when the phase changes.

When directly deposited, spiral inductors, such as shown in FIG. 12, areadvantageous functionally; however, other inductors may be used. Theinductor is electrically connected to the upper leads of theinterdigital electrodes to form the sensor 16, as shown in FIG. 24. InFIG. 24, the inductor 240 is embedded with the capacitor. Inenvironments where the cured material and capacitive elements arepartially encased in metal or other encasements which reduce thetransmissivity of radio frequency energy, the inductor can be mountedexternal to the encasement and connected to the capacitive element(e.g., flex circuits). This is illustrated in FIG. 25. FIG. 26illustrates the sensor embedded in a rectangular cube.

Another means of developing the capacitive element for wear measurementis to use interdigital electroplates, as shown in FIG. 27. Similar tothe device shown in FIG. 22, the metal used for the electrodes is ametal that can wear away more easily than the surface of which wear isto be measured. The capacitive device is placed while the volume isliquid. The liquid is cured to a solid, thereby embedding the capactivedevice. As the surface wears away, the area of the electric plates wearsaway, altering the capacitance of the device.

FIG. 28 illustrates an embodiment of the interdigital electroplates withtemperature sensitive dielectric, thermomagnetic or a phase transitiondielectric 280 between the electroplates. The temperature sensitivedielectric or the phase transition dielectric 280 add their respectivefunctionality as described above.

Another embodiment utilizes direct deposition of one or moreinterdigital electrodes as a thin film positioned such that theelectrodes are parallel to the surface of wear. The electrodes arepositioned along an outer surface of the material for which wear is tobe determined. If the electrodes are coated with a layer temperaturesensitive dielectric, thermomagnetic or a phase transition material; theembodiment can be used for both wear and thermal measurements.

For wear measurement, an inductor is placed on the upper surface of amaterial whose thickness is lessen with wear and the lower surface is incontact with a conductive material. As material wears, inductancedecreases due to increased proximity to conductive material. If theinterdigital electrodes are used and are coated with a layer temperaturesensitive dielectric, thermomagnetic or a phase transition material; theembodiment can be used for both wear and thermal measurements.

EXAMPLE 4 Sensor for Displacement Measurements

A first embodiment of a sensor for displacement measurements isillustrated in FIG. 31. This embodiment comprises two parallelelectroplates 290 (negative) and 291 (positive). One electroplate isstationary. The other electroplate has an opposite charge and movesperpendicular to its surface. The direction of electric field E isindicated. The capacitance, C(x), is dependent upon the distance thatthe plates are separated, x. $\begin{matrix}{{C(x)} = \frac{ɛ_{0}{lw}}{x}} & (30)\end{matrix}$When the electroplate capacitor is coupled to an inductor, such as thesquare spiral inductor illustrated in FIG. 12, thus forming a parallelcircuit, the resonant electrical frequency of the circuit is$\begin{matrix}{{\omega(x)} = {\frac{1}{\sqrt{{LC}(x)}}.}} & (31)\end{matrix}$Although a square spiral is illustrated in FIG. 12, other inductordesigns can be used. The complete sensing element, showing inductor 300,is illustrated in FIG. 30.

Inclusion of the equation for capacitance, Equation (30), into that forresonant frequency, Equation (31), results in the following expressionwhich relates the resonant frequency to plate separation distance.$\begin{matrix}{{\omega(x)} = \left\lbrack \frac{L\quad ɛ_{0}{lw}}{x} \right\rbrack^{{- 1}/2}} & (32)\end{matrix}$The variation in frequency with respect to separation plate separationdistance is $\begin{matrix}{\frac{\mathbb{d}{\omega(x)}}{\mathbb{d}x} = {+ {{{\frac{1}{2}\left\lbrack \frac{L\quad ɛ_{0}{lw}}{x} \right\rbrack}^{{- 3}/2}\left\lbrack \frac{L\quad ɛ_{0}{lw}}{x^{2}} \right\rbrack}.}}} & (33)\end{matrix}$The frequency variation is dominated by the inverse quadratic term. Thefrequency will change more pronouncely as the plates are brought closertogether. The sensitivity of the frequency with respect to theseparation distance is of order x^(−1/2). Capacitance variation withdisplacement is shown in FIGS. 31 and 32. FIG. 31 shows results of atotal displacement of 0.10 inches using displacement increments of 0.025in. A more refined resolution is shown in FIG. 32, where increments of0.005 in were used for a total displacement of 0.025 inches. Thedielectric is ambient air.

Key design parameters of this embodiment are the total length ofelectroplates, l, and the width of the plates, w. The equations shown inTable XII relate the sensitivity of the resonant frequency changes tochanges in the aforementioned key parameters (i.e., sensitivity changesresulting from a variation in a parameter). TABLE XII Sensitivityresulting from parameter change Parameter Sensitivity Length ofelectroplates$\frac{\mathbb{d}\omega}{\mathbb{d}l} = {- {{\frac{1}{2}\left\lbrack \frac{L\quad ɛ_{0}{lw}}{x} \right\rbrack}^{{- 3}/2}\left\lbrack \frac{L\quad ɛ_{0}w}{x} \right\rbrack}}$Width of electroplates$\frac{\mathbb{d}\omega}{\mathbb{d}w} = {- {{\frac{1}{2}\left\lbrack \frac{L\quad ɛ_{0}{lw}}{x} \right\rbrack}^{{- 3}/2}\left\lbrack \frac{L\quad ɛ_{0}l}{x} \right\rbrack}}$

As the plates 290 and 291 are made longer or wider, the resonantfrequency becomes less sensitive to displacement, as can be seen fromthe two sensitivity expressions.

A second embodiment, illustrated in FIG. 33, comprises a dielectric 330of thickness, b, affixed to a stationary electroplate 331 (positive).The voltage across the electroplates 331 and 332 (negative) is dependentupon the electric field through the dielectric, E_(b), and the free air,E. $\begin{matrix}\begin{matrix}{V = {- {\int_{0}^{x}{{E(l)} \cdot {\mathbb{d}l}}}}} \\{= {- {\int_{0}^{x}{{E(l)}{\cos\left( {180{^\circ}} \right)}{\mathbb{d}l}}}}} \\{= {\int_{o}^{x}{{E(l)}{\mathbb{d}l}}}} \\{= {{E_{b}b} + {E\left( {x - b} \right)}}}\end{matrix} & (34)\end{matrix}$The electric field in the dielectric is $\begin{matrix}{E_{b} = {\frac{E}{\kappa}.}} & (35)\end{matrix}$Therefore the voltage across the plates 331 and 332 is $\begin{matrix}\begin{matrix}{V = {E\left( {\frac{b}{\kappa} + x - b} \right)}} \\{= {E\left\lbrack {x - {b\left( {1 - \frac{1}{\kappa}} \right)}} \right\rbrack}}\end{matrix} & (36)\end{matrix}$The capacitance across the plates 331 and 332 is $\begin{matrix}\begin{matrix}{{C(x)} = \frac{q}{V}} \\{= \frac{ɛ_{0}{wlE}}{E\left( {x - {b\left( {1 - \frac{1}{\kappa}} \right)}} \right\rbrack}} \\{= \frac{ɛ_{0}{wl}}{\left\lbrack {x - {b\left( {1 - \frac{1}{\kappa}} \right)}} \right\rbrack}}\end{matrix} & (37)\end{matrix}$resulting in the following expression for resonant frequency$\begin{matrix}{{\omega(x)} = {\left\lbrack \frac{L\quad ɛ_{0}{lw}}{\left\lbrack {x - {b\left( {1 - \frac{1}{\kappa}} \right)}} \right\rbrack} \right\rbrack^{{- 1}/2}.}} & (38)\end{matrix}$Equation (38) is the more general expression for the displacement sensorembodied as capacitive plates that have relative translations that areperpendicular to each other. When no dielectric is present, it reducesto that of Equation (31).

The variation in frequency with respect to separation plate separationdistance is $\begin{matrix}{\frac{\mathbb{d}{\omega(x)}}{\mathbb{d}x} = {{+ {\frac{1}{2}\left\lbrack \frac{L\quad ɛ_{0}{lw}}{\left\lbrack {x - {b\left( {1 - \frac{1}{\kappa}} \right)}} \right\rbrack} \right\rbrack}^{{- 3}/2}}{\frac{L\quad ɛ_{0}{lw}}{\left\lbrack {x - {b\left( {1 - \frac{1}{\kappa}} \right)}} \right\rbrack^{2}}.}}} & (39)\end{matrix}$The frequency variation is dominated by the inverse quadratic term. Thefrequency will change more pronouncely as the plates are brought closertogether. The sensitivity of the frequency with respect to theseparation distance is of order$\left\lbrack {x - {b\left( {1 - \frac{1}{\kappa}} \right)}} \right\rbrack^{{- 1}/2}.$Sensitivity is more pronounced for dielectrics of either increasedthickness or higher dielectric constant.

A third embodiment, shown in FIG. 34, comprises two parallelelectroplates 370 (negative) and 371 (positive). One electroplate isstationary. The other electroplate has an opposite charge and movesparallel to its surface. The capacitance, C(x), is dependent upon thelength, x, that the plates overlap. The effective area of the capacitoris dependent upon the plates' overlap. The plates are separated by adistance, d. Each plate has width w. The resulting capacitance is$\begin{matrix}{{C(x)} = {\frac{\kappa\quad ɛ_{0}{wx}}{d}.}} & (40)\end{matrix}$When the electroplate capacitor is coupled to an inductor, as shown inFIG. 32, thus forming a parallel circuit, the resonant electricalfrequency of the circuit is provided by Equation (31). Although a squarespiral is shown in FIG. 12, other inductor designs can be used. Thecomplete sensor is shown in FIG. 34. Inclusion of the equation forcapacitance (Equation (41) into that for resonant frequency (Equation(31)) results in the following expression which relates the resonantfrequency to plate separation distance $\begin{matrix}{{\omega(x)} = {\left\lbrack \frac{L\quad\kappa\quad ɛ_{0}{wx}}{d} \right\rbrack^{{- 1}/2}.}} & (41)\end{matrix}$The variation in frequency with respect to the plate overlap length, x,is $\begin{matrix}{\frac{\mathbb{d}{\omega(x)}}{\mathbb{d}x} = {+ {{{\frac{1}{2}\left\lbrack \frac{L\quad\kappa\quad ɛ_{0}{xw}}{d} \right\rbrack}^{{- 3}/2}\left\lbrack \frac{L\quad\kappa\quad ɛ_{0}w}{d} \right\rbrack}.}}} & (42)\end{matrix}$The sensitivity of the frequency with respect to the separation distanceis of order x^(−3/2). Capacitance variation with displacement is shownin FIG. 35. FIG. 35 illustrates results of a total displacement of 0.475inches using displacement increments of 0.025 in. The dielectric isambient air.

Key design parameters of this embodiment are width of the plates, w;separation of plates, d, and the dielectric constant, κ. The equationsin Table XIII relate the sensitivity of the resonant frequency changesto changes in the aforementioned key parameters (i.e., sensitivitychange resulting from parameter variation) TABLE XIII Sensitivityresulting from parameter change Parameter Sensitivity Separationdistance$\frac{\mathbb{d}{\omega(x)}}{\mathbb{d}(d)} = {+ {{\frac{1}{2}\left\lbrack \frac{L\quad{\kappa ɛ}_{0}{wx}}{d} \right\rbrack}^{{- 3}/2}\left\lbrack \frac{L\quad{\kappa ɛ}_{0}{wx}}{d^{2}} \right\rbrack}}$Width of electroplates$\frac{\mathbb{d}{\omega(x)}}{\mathbb{d}w} = {- {{\frac{1}{2}\left\lbrack \frac{L\quad{\kappa ɛ}_{0}{wx}}{d} \right\rbrack}^{{- 3}/2}\left\lbrack \frac{L\quad{\kappa ɛ}_{0}x}{d} \right\rbrack}}$Dielectric constant$\frac{\mathbb{d}{\omega(x)}}{\mathbb{d}\kappa} = {- {{\frac{1}{2}\left\lbrack \frac{L\quad{\kappa ɛ}_{0}{wx}}{d} \right\rbrack}^{{- 3}/2}\left\lbrack \frac{L\quad{\kappa ɛ}_{0}x}{d} \right\rbrack}}$

As the plates are made wider or if a larger dielectric constant is used,the resonant frequency becomes less sensitive to displacement, as can beseen from the two sensitivity expressions. Decreasing the separationdistance of the plates increases the sensitivity to displacement.

A fourth embodiment is illustrated in FIG. 36. If a structural member360 (rod, truss, beam, etc) of known elastic modulus, E, and crosssectional area, A, has two rigid electrically capacitive plates 361affixed to it (either externally or embedded), axial compression ortension can be measured. The plates must be oriented such that the axialforce, P, is perpendicular to their surface. The axial load isP=εAE   (43)The elongation per unit length or strain, ε, is determined by$\begin{matrix}{ɛ = {\frac{l - l_{0}}{l_{0}} = {\frac{\Delta\quad l}{l_{0}}.}}} & (44)\end{matrix}$The capacitance, C, is given by $\begin{matrix}{C = {ɛ_{c}\frac{A_{c}}{d}}} & (45)\end{matrix}$where ε_(c), A_(c) and d are the permittivity, capacitor plate area andplate separation, respectively. Any change in capacitance is dependentupon the elongation of the member. Hence, $\begin{matrix}{{\Delta\quad l} = {{l - l_{0}} = {{\Delta\quad d} = {ɛ_{c}{A_{c}\left( {\frac{1}{C_{2}} - \frac{1}{C_{1}}} \right)}}}}} & (46)\end{matrix}$Therefore, any applied axial load is $\begin{matrix}{P = {\frac{{AE}\quad ɛ_{c}A_{c}}{l_{0}}\left( {\frac{1}{C_{2}} - \frac{1}{C_{1}}} \right)}} & (47)\end{matrix}$This embodiment allows axial load to be determined by measuring changesin capacitance. When the capacitor is electrically coupled to aninductor, axial load is now determined by changes in measured resonantfrequency.

Interdigital electrodes could be used in lieu of the capacitive plates.A resistive sensor is bonded to a surface for which it is sensing shear.The surface material, the bond adhesive and the resistor all havedifferent moduli of elasticity. When strained, each deforms separately.The effect is minimized when the substrate modulus is far higher thatthe adhesive and resistive material. However, for materials with lowmodulus, the resistive material could significantly dominate the overallcomposite modulus due to all constituent layers. Use of the capacitoreliminates this effect because the electroplates (electrodes) can moveindependent of each other.

EXAMPLE 5 Sensor for Fluid Level Measurement

A first embodiment of a fluid level sensor, illustrated in FIG. 29,comprises two parallel electroplates 290 and 291. The direction ofelectric field E is indicated. In FIG. 37, a dielectric medium (κ, otherthan air) 370 fills a portion of the gap between the electric plates,which would alter the capacitance in a manner similar to having thecapacitor partially immersed in the medium. The capacitance, C(x), isdependent upon the immersion depth, x, and is the combination of thecapacitance of that portion of the electroplate that is not immersed inthe medium and the capacitance of the portion that is immersed in themedium. The two portions of the capacitor act as a parallel capacitorsince they share the same electric field. $\begin{matrix}\begin{matrix}{{C(x)} = {C_{free} + C_{immersed}}} \\{= {{\left( {l - x} \right)\frac{ɛ_{0}w}{d}} + {\frac{ɛ_{0}w}{d}\kappa\quad x}}} \\{= {\frac{ɛ_{0}w}{d}\left\lbrack {l + {x\left( {\kappa - 1} \right)}} \right\rbrack}}\end{matrix} & (48)\end{matrix}$When the capacitor is not immersed (i.e., dielectric medium level, x=0),the capacitance is $\begin{matrix}{{C(x)} = \frac{ɛ_{0}w\quad l}{d}} & (49)\end{matrix}$The capacitor completely immersed (i.e., dielectric medium level, x=l)has capacitance of $\begin{matrix}{{C(x)} = \frac{\kappa\quad ɛ_{0}\omega\quad l}{d}} & (50)\end{matrix}$

When the electroplate capacitor is coupled to an inductor, such as thesquare spiral inductor illustrated in FIG. 12, thus forming a parallelcircuit, the resonant electrical frequency of the circuit is$\begin{matrix}{\omega = {\frac{1}{\sqrt{{LC}(x)}}.}} & (51)\end{matrix}$Although a square spiral is shown in FIG. 12, other inductor designs canbe used. The complete sensor is shown in FIG. 38.

Inclusion of the equation for capacitance (Equation (48) into that forresonant frequency (Equation (51)) results in the following expressionwhich relates the resonant frequency to immersion depth $\begin{matrix}{\omega = {\left\lbrack {\frac{L\quad ɛ_{0}w}{d}\left\lbrack {l + {x\left( {\kappa - 1} \right)}} \right\rbrack} \right\rbrack^{{- 1}/2}.}} & (52)\end{matrix}$

Key design parameters of this embodiment are the total length ofelectroplates, l; width of the plates, w; separation of the plates, d;and the dielectric constant, κ, of the medium in which the plates areimmersed. The equations shown in Table XIV relate the sensitivity of theresonant frequency changes to changes in the aforementioned parameters(i.e., sensitivity changes resulting from parameter variation). TABLEXIV Sensitivity resulting from parameter changes Parameter SensitivityLength of electroplates$\frac{\mathbb{d}\omega}{\mathbb{d}l} = {- {{\frac{1}{2}\left\lbrack {\frac{L\quad ɛ_{0}w}{d}\left\lbrack {l + {x\left( {\kappa - 1} \right)}} \right\rbrack} \right\rbrack}^{{- 3}/2}\left\lbrack \frac{L\quad ɛ_{0}w}{d} \right\rbrack}}$Width of electroplates$\frac{\mathbb{d}\omega}{\mathbb{d}w} = {- {{\frac{1}{2}\left\lbrack {\frac{L\quad ɛ_{0}w}{d}\left\lbrack {l + {x\left( {\kappa - 1} \right)}} \right\rbrack} \right\rbrack}^{{- 3}/2}\left\lbrack \frac{L\quad{ɛ_{0}\left\lbrack {l + {x\left( {\kappa - 1} \right)}} \right\rbrack}}{d} \right\rbrack}}$Separation of electroplates$\frac{\mathbb{d}\omega}{\mathbb{d}(d)} = {+ {{\frac{1}{2}\left\lbrack {\frac{L\quad ɛ_{0}w}{d}\left\lbrack {l + {x\left( {\kappa - 1} \right)}} \right\rbrack} \right\rbrack}^{{- 3}/2}\left\lbrack \frac{L\quad ɛ_{0}{w\left\lbrack {l + {x\left( {\kappa - 1} \right)}} \right\rbrack}}{d^{2}} \right\rbrack}}$Dielectric constant$\frac{\mathbb{d}\omega}{\mathbb{d}\kappa} = {- {{\frac{1}{2}\left\lbrack {\frac{L\quad ɛ_{0}w}{d}\left\lbrack {l + {x\left( {\kappa - 1} \right)}} \right\rbrack} \right\rbrack}^{{- 3}/2}\left\lbrack \frac{L\quad ɛ_{0}{wx}\quad\kappa}{d} \right\rbrack}}$

As the plates are made longer or wider, the resonant frequency becomesless sensitive to changes in dielectric constant level, as can be seenfrom the first two sensitivity expressions. The resonant frequencysensitivity to plate separation is inversely quadratic, which results inthe sensitivity changing quadratically as the plates are placed closertogether. When the electroplate capacitor is to be used for viscousfluids, the plate separation also effects any capillary action of thefluid.

A consideration for using the sensor for viscous fluids is the effect ofresidual fluid film on the electroplates after the plates have beenremoved from the fluid. Many dielectrics leave a film residue whenremoved from the electroplates. FIG. 39 illustrates electroplates with aseparation distance, d. A film of thickness, b, is to the left of eachplate. The separation of the plates is far greater than the thickness ofthe film (i.e., b <<d). The voltage across the electroplates isdependent upon the electric field through the dielectric, E_(b), and thefree air, E. $\begin{matrix}\begin{matrix}{V = {- {\int_{0}^{d}{E \cdot {\mathbb{d}l}}}}} \\{= {- {\int_{0}^{d}{E\quad{\cos\left( {180{^\circ}} \right)}{\mathbb{d}l}}}}} \\{= {\int_{0}^{d}{E{\mathbb{d}l}}}} \\{= {{2E_{b}b} + {E\left( {d - {2b}} \right)}}}\end{matrix} & (53)\end{matrix}$The electric field in the dielectric is provided by Equation (35).Therefore the voltage across the plates is $\begin{matrix}\begin{matrix}{V = {E\left( {\frac{2b}{\kappa} + d - {2b}} \right)}} \\{= {E\left( {d - {2{b\left( {1 - \frac{1}{\kappa}} \right)}}} \right)}}\end{matrix} & (54)\end{matrix}$To determine the effect of the dielectric, it is necessary to examinethe term $\left( {1 - \frac{1}{\kappa}} \right)$for extreme values of κ. The lower bounds of value that the dielectriccan have is the value in vacuum (κ=1). The dielectric value of air isapproximately 1 (K≈1). Therefore if no dielectric film was present,$\begin{matrix}{\left( {1 - \frac{1}{\kappa}} \right) \approx 0} & (55)\end{matrix}$For cases in which the dielectric constant is greater than 1,$\begin{matrix}{{\lim_{K->\infty}\left( {1 - \frac{1}{\kappa}} \right)} = 1.} & (56)\end{matrix}$Therefore $\begin{matrix}{0 \leq \left( {1 - \frac{1}{\kappa}} \right) \leq 1.} & (57)\end{matrix}$which results in the following two voltage extrema $\begin{matrix}{{Ed} \leq {E\left( {d - {2{b\left( {1 - \frac{1}{\kappa}} \right)}}} \right)} \leq {{E\left( {d - {2b}} \right)}.}} & (58)\end{matrix}$The latter extrema is that which can be used to determine the effect ofthe dielectric film. Using the latter extrema, the voltage across theelectroplates isV=E(d−2b)≅Ed for b<<d.   (59)Thus, the film has a negligible effect on the voltage across theelectroplates and, thus, the capacitance across the plates.

A second embodiment of a fluid level sensor, illustrated in FIG. 40,comprises n pairs of parallel electroplates. A pair is any surface of apositive plate 400 facing the surface of a negative plate 401. The keygeometric parameters are those provided in FIG. 29. The direction ofelectric field E is indicated. A dielectric medium 370 (κ, other thanair) fills a portion of the gap between the electric plates 400 and 401that would alter the capacitance in a manner similar to having theplates 400 and 401 partially immersed in the medium. The capacitance,C(x), is provided by Equations (48), (49) and (50), each multiplied byn. The resonant electrical frequency is provided by Equation (52).Although a square spiral is shown in FIG. 12, other inductor designs canbe used. The complete sensor is shown in FIG. 41. Inclusion of theequation for capacitance (Equation (48)) into that for resonantfrequency (Equation (51)), modified by factor n, results in thefollowing expression which relates the resonant frequency to immersiondepth $\begin{matrix}{\omega = \left\lbrack {\frac{n\quad L\quad ɛ_{0}w}{d}\left\lbrack {l + {x\left( {\kappa - 1} \right)}} \right\rbrack} \right\rbrack^{{- 1}/2}} & (60)\end{matrix}$The expression for resonant frequency is that of the single set ofelectroplates with a multiplying factor, n. Hence multiple plates can beused to tailor the resonant frequency so that its variation is within aspecified range.

Key design parameters of this embodiment are number of parallelelectroplate sets, n; total length of electroplates, l; width of theplates, w; separation of the plates, d, and the dielectric constant, κ,of the medium in which the plates are immersed. The equations in TableXV relate the sensitivity of the resonant frequency changes to changesin the aforementioned key parameters (i.e., sensitivity changesresulting from parameter variation). TABLE XV Sensitivity variationresulting from parameter change Parameter Sensitivity Number ofelectroplate sets$\frac{\mathbb{d}\omega}{\mathbb{d}n} = {- {{\frac{1}{2}\left\lbrack {\frac{{nL}\quad ɛ_{0}w}{d}\left\lbrack {l + {x\left( {\kappa - 1} \right)}} \right\rbrack} \right\rbrack}^{{- 3}/2}\left\lbrack \frac{L\quad ɛ_{0}{w\left( {l + {x\left( {\kappa - 1} \right)}} \right)}}{d} \right\rbrack}}$Length of electroplates$\frac{\mathbb{d}\omega}{\mathbb{d}l} = {- {{\frac{1}{2}\left\lbrack {\frac{{nL}\quad ɛ_{0}w}{d}\left\lbrack {l + {x\left( {\kappa - 1} \right)}} \right\rbrack} \right\rbrack}^{{- 3}/2}\left\lbrack \frac{{nL}\quad ɛ_{0}w}{d} \right\rbrack}}$Width of electroplates$\frac{\mathbb{d}\omega}{\mathbb{d}w} = {- {{\frac{1}{2}\left\lbrack {\frac{{nL}\quad ɛ_{0}w}{d}\left\lbrack {l + {x\left( {\kappa - 1} \right)}} \right\rbrack} \right\rbrack}^{{- 3}/2}\left\lbrack \frac{{nL}\quad{ɛ_{0}\left( {l + {x\left( {\kappa - 1} \right)}} \right)}}{d} \right\rbrack}}$Separation of electroplates$\frac{\mathbb{d}\omega}{\mathbb{d}(d)} = {+ {{\frac{1}{2}\left\lbrack {\frac{{nL}\quad ɛ_{0}w}{d}\left\lbrack {l + {x\left( {\kappa - 1} \right)}} \right\rbrack} \right\rbrack}^{{- 3}/2}\left\lbrack \frac{{nL}\quad ɛ_{0}{w\left( {l + {x\left( {\kappa - 1} \right)}} \right)}}{d^{2}} \right\rbrack}}$Dielectric constant$\frac{\mathbb{d}\omega}{\mathbb{d}\kappa} = {- {{\frac{1}{2}\left\lbrack {\frac{{nL}\quad ɛ_{0}w}{d}\left\lbrack {l + {x\left( {\kappa - 1} \right)}} \right\rbrack} \right\rbrack}^{{- 3}/2}\left\lbrack \frac{{nL}\quad ɛ_{0}{wx}}{d} \right\rbrack}}$

As the plates are made longer or wider, the resonant frequency becomesless sensitive to changes in dielectric constant level, as can be seenfrom the second and third sensitivity expressions. The resonantfrequency sensitivity to plate separation is inversely quadratic, whichresults in the sensitivity changing quadratically as the plates areplaced closer together. When the electroplate capacitor is to be usedfor viscous fluids, the plate separation also effects any capillaryaction of the fluid. Increasing the number of electroplate setsincreases the effect of the other key parameters of resonant frequencysensitivity. Therefore, more sensitivity is achieved when multiple platesets are used and the separation distance between plates of oppositecharge is small. However, as the other parameters are increased, thesensitivity is decreased. Another consideration for using the sensor forviscous fluids is the effect of residual fluid film on the electroplatesafter the plates have been removed from the fluid.

A third embodiment, illustrated in FIG. 42, comprises n pair of parallelinterdigital electrodes 420 for the capacitor. The advantage of thismethod is that the entire sensor can be embodied as a lamina (e.g., thinfilm). The fluid sensor can be directly deposited to the wall of anon-conductive container via direct deposition. A pair is any positiveelectrode that neighbors a negative electrode. A cross sectional A-A′ ofelectrically charged interdigital capacitor with electric fieldillustrated is FIG. 43. The electric field 431 starts from the positiveelectrodes 430 and ends at the negative electrodes 432, shown onsubstrate 433. Unlike the first and second embodiments, the electricfield is not perpendicular to the electrodes. Portions of the field nearthe electrodes are parallel to the plane of the electrodes. The electrichas its highest concentration near the plane of the electrodes.

In FIG. 44, a dielectric medium (κ) 440 is in contact with m pairs ofelectrodes 441 (e.g., placed in a fluid such that m electrode pairs aresubmersed). The capacitance, C(m), is dependent upon the number ofelectrode pairs, m, in contact with the dielectric and those pairs whichare not in contact. All of the electrode pairs 441 are parallelcapacitors since they share the same electric field. Another featurethat distinguishes this embodiment from the first and second is that themeasurement variation to dielectric contact are discrete with theinterdigital electrodes, but are continuous when the electroplates areused. $\begin{matrix}\begin{matrix}{{C(m)} = {C_{free} + C_{immersed}}} \\{= {{\left( {n - m} \right)\frac{ɛ_{0}w\quad l}{d}} + \frac{m\quad\kappa\quad ɛ_{0}w\quad l}{d}}} \\{= {\frac{ɛ_{0}{wl}}{d}\left\lbrack {n + {m\left( {\kappa - 1} \right)}} \right\rbrack}}\end{matrix} & (61)\end{matrix}$When the capacitor is not immersed (i.e., dielectric medium level, m=0),the capacitance is $\begin{matrix}{{C(m)} = \frac{ɛ_{0}{nw}\quad l}{d}} & (62)\end{matrix}$The capacitor completely immersed (i.e., dielectric medium level, m=n)has capacitance of $\begin{matrix}{{C(m)} = \frac{\kappa\quad ɛ_{0}n\quad\omega\quad l}{d}} & (63)\end{matrix}$When the electroplate capacitor is coupled to an inductor, such as thesquare spiral illustrated in FIG. 12, thus forming a parallel circuit,the resonant electrical frequency of the circuit is $\begin{matrix}{\omega = {\frac{1}{\sqrt{{LC}(m)}}.}} & (64)\end{matrix}$Although a square spiral is shown in FIG. 12, other inductor designs canbe used. The complete sensor is shown in FIG. 44.

Inclusion of the equation for capacitance (Equation (61) into that forresonant frequency (Equation (64)) results in the following expressionwhich relates the resonant frequency to immersion depth $\begin{matrix}{\omega = {\left\lbrack {\frac{L\quad ɛ_{0}w\quad l}{d}\left\lbrack {n + {m\left( {\kappa - 1} \right)}} \right\rbrack} \right\rbrack^{{- 1}/2}.}} & (65)\end{matrix}$

Key design parameters of this embodiment are number of parallelelectrode pairs, n; length of positive and negative electrode overlap,l; width of the electrodes, w; separation of the electrodes, d, and thedielectric constant, κ, of the medium in which the electrodes areimmersed. The equations in Table XVI relate the sensitivity of theresonant frequency changes to changes in the aforementioned keyparameters (i.e., sensitivity changes resulting from parameter change).TABLE XVI Sensitivity resulting from parameter change ParameterSensitivity Number of interdigital electrode pairs$\frac{\mathbb{d}\omega}{\mathbb{d}n} = {- {{\frac{1}{2}\left\lbrack {\frac{L\quad ɛ_{0}{wl}}{d}\left\lbrack {n + {m\left( {\kappa - 1} \right)}} \right\rbrack} \right\rbrack}^{{- 3}/2}\left\lbrack \frac{L\quad ɛ_{0}{wl}}{d} \right\rbrack}}$Length of positive and negative electrode overlap$\frac{\mathbb{d}\omega}{\mathbb{d}l} = {- {{\frac{1}{2}\left\lbrack {\frac{L\quad ɛ_{0}{wl}}{d}\left\lbrack {n + {m\left( {\kappa - 1} \right)}} \right\rbrack} \right\rbrack}^{{- 3}/2}\left\lbrack \frac{L\quad ɛ_{0}{w\left\lbrack {n + {m\left( {\kappa - 1} \right)}} \right\rbrack}}{d} \right\rbrack}}$Width of electrodes$\frac{\mathbb{d}\omega}{\mathbb{d}w} = {- {{\frac{1}{2}\left\lbrack {\frac{L\quad ɛ_{0}{wl}}{d}\left\lbrack {n + {m\left( {\kappa - 1} \right)}} \right\rbrack} \right\rbrack}^{{- 3}/2}\left\lbrack \frac{L\quad ɛ_{0}{l\left\lbrack {n + {m\left( {\kappa - 1} \right)}} \right\rbrack}}{d} \right\rbrack}}$Separation of electrodes$\frac{\mathbb{d}\omega}{\mathbb{d}(d)} = {+ {{\frac{1}{2}\left\lbrack {\frac{L\quad ɛ_{0}{wl}}{d}\left\lbrack {n + {m\left( {\kappa - 1} \right)}} \right\rbrack} \right\rbrack}^{{- 3}/2}\left\lbrack \frac{L\quad ɛ_{0}{{wl}\left\lbrack {n + {m\left( {\kappa - 1} \right)}} \right\rbrack}}{d^{2}} \right\rbrack}}$Dielectric constant$\frac{\mathbb{d}\omega}{\mathbb{d}\kappa} = {- {{\frac{1}{2}\left\lbrack {\frac{L\quad ɛ_{0}{wl}}{d}\left\lbrack {n + {m\left( {\kappa - 1} \right)}} \right\rbrack} \right\rbrack}^{{- 3}/2}\left\lbrack \frac{{mL}\quad ɛ_{0}{wl}}{d} \right\rbrack}}$

As the electrode overlap becomes longer or as the electrodes are madewider, the resonant frequency becomes less sensitive to changes indielectric constant level, as can be seen from the second and thirdsensitivity expressions. The resonant frequency sensitivity to electrodeseparation is inversely quadratic, which results in the sensitivitychanging quadratically as the plates are placed closer together.Increasing the number of electrode pairs increases the sensitivityeffect of the other key parameters. Therefore, more sensitivity isachieved when multiple electrode pairs are used and the separationdistance between plates of opposite charge is reduced. However, as theother parameters are increased, the sensitivity due to more electrodepairs is decreased.

Another consideration for using the sensor for viscous fluids is theeffect of residual fluid film on the electroplates after the plates havebeen removed from the fluid.

The effect of dielectric film on the interdigital electrodes is morepronounced than on the electroplates. FIG. 45 illustrates some of theelectrodes with a thin residual film. The electric field E is alsoindicated. The field lines are nearly parallel to the film, resulting inthe electrical field being exposed to the dielectric at the part of thefield that has the highest strength (near the surface). As a result ofthe more pronounced effect of residual film, the interdigital electrodesare suitable for viscous fluids (e.g., water, gas, alcohol).

As an experimental example, a magnetic field response fluid-level sensorembodiment is shown in FIG. 38. The sensor consists of two capacitiveplates electrically coupled to an inductor. The capacitor was placed ina cylindrical container while the inductor remained outside thecontainer. The container was filled with hydraulic fluid. As the fluidfilled the void between the plates, the effective dielectric increasedproportional to fluid immersion, thus changing the sensor's resonantfrequency. Frequency measurements for two 9-inch fluid-level sensors ofdifferent widths are shown in FIG. 46. As the levels increased, thefrequencies decreased. Fluid level was increased using 0.5 inincrements. A fluid-level of 9 inches resulted in frequency reductionsof over 1.1 MHz (⅛ in plate width) and 0.8 MHz ({fraction (1/16)} inplate width) from that of the empty container. The sensor embodimentillustrated in FIG. 21 can also be used for measuring the fluid levelsof non-viscous fluids. The electric field of the interdigital electrodesarcs from one positive electrode to its neighboring negative electrode.Most of the interdigital electrode's electric field is near the plane ofthe electrodes, whereas the electric field of the capacitive plates isperpendicular to the plates. The interdigital electrode's electric fieldhas proportionally more exposure to the viscous fluid film residue (andmore dielectric exposure) than that of the plates. The capacitive platesare necessary when viscous fluids are used because any residual film hasa negligible effect on measurements. The amount of plate separation isdesigned to eliminate capillary effects. When non-viscous fluids areused (e.g., water, gasoline, alcohol), the interdigital electrodes donot require the volume necessary for plates since they can be placed onthin-film dielectrics or directly deposited to a surface. Theinterdigital electrodes are easier to fabricate and mount.

EXAMPLE 6 Sensor for Contact Measurement

A first embodiment of a sensor for contact measurement uses two separatecomponents affixed to either surface. A component can either be a L-Ccircuit, inductor or capacitor. Table XVII lists combinations ofcomponents and their responses before and after contact. In (1) and (2),the circuit is altered by changing the value of existing constituents(e.g., adding capacitance or inductance). A circuit is created in (3)when the surfaces contact.

In a second embodiment, an L-C circuit is shorted when contact is made.(1) or (2) are the desired combinations. Magnetic field responses existbefore and after contact. Hence, contact is gauged by a shift infrequency response. In the other cases, the response either existsbefore or after contact but not both.

Measuring the bond between two surfaces can be interrogated in themanner similar to contact. Component combinations of (1)-(4) can be usedto determine bond also. The method can be extended to determine degreeof separation using the numeric encoding method outlined in Tables VIIIand Table X. The electrical contacts are distributed in an arraythroughout a first surface. The surface array has an inductor andcapacitor which allows it to resonate (frequency is the resultant ofsingle inductor and capacitor) even when not in contact with the othersurface. A mating array of capacitors is on a second surface with theirelectrical leads facing toward and beneath those of the array on thesecond surface. When both surfaces are bonded, the resonant is theresultant of all the capacitors and a single inductor. If contact(hence, bond) is severed, the resonant will shift in frequency. As morecontacts are broken, the frequency increases. TABLE XVIII ContactCombinations Component First Second Response prior combinationscomponent component to contact Response after contact (1) L-C circuitCapacitor $\omega = \frac{1}{\sqrt{LC}}$$\omega = \frac{1}{\sqrt{2{LC}}}$ (2) L-C circuit Inductor$\omega = \frac{1}{\sqrt{LC}}$ $\omega = \sqrt{\frac{2}{LC}}$ (3)Inductor Capacitor Does not exist (Circuit is not complete)$\omega = \frac{1}{\sqrt{LC}}$ (4) L-C circuit Conductive surface$\omega = \frac{1}{\sqrt{LC}}$ Does not exist (Circuit is shorted whenit contacts conductiove surface)

Although the invention has been described relative to specificembodiments thereof, there are numerous variations and modificationsthat will be readily apparent to those skilled in the art in light ofthe above teachings. It is therefore to be understood that, within thescope of the appended claims, the invention may be practiced other thanas specifically described.

1. A magnetic field response measurement acquisition system, comprising:one or more inductively powered magnetic field response sensors; antennameans for transmitting radio frequency energy to and receiving radiofrequency energy from said one or more sensors; an interrogation meansfor regulating said radio frequency transmission and reception, and foranalyzing the signals received from said one or more sensors, whereinsaid processor means can interrogate multiple sensors concurrently usinga single acquisition channel, does not require that said signals fromsaid one or more sensors be transmitted as modulated signals on a radiofrequency carrier, and can acquire more than one measurement from eachsaid sensor.
 2. The acquisition system of claim 1, wherein said antennameans is a single switching antenna.
 3. The acquisition system of claim1, wherein said antenna means is separate transmission and receivingantennae.
 4. The acquisition system of claim 1, wherein saidinterrogation means is portable.
 5. The acquisition system of claim 1,wherein said interrogation means is handheld.
 6. The acquisition systemof claim 1, wherein said one or more sensors are selected from the groupconsisting of inductor-capacitor circuits powered by Faraday inductionand inductor-capacitor-resistor circuits powered by Faraday induction.7. The acquisition system of claim 6, wherein the responses of saidsensors change corresponding with a change in the physical states thatsaid sensors measure.
 8. The acquisition system of claim 7, wherein theattributes of said responses are one or more attributes selected fromthe group consisting of frequency, amplitude and bandwidth.
 9. Theacquisition system of claim 1, wherein said system has the ability toacquire measurements resulting from changes in capacitor geometric,capacitor dielectric, inductor geometric, inductor permeability, andresistance.
 10. The acquisition system of claim 1, wherein said antennameans are one or more broadband antennas.
 11. The acquisition system ofclaim 1 having more than one sensor, wherein the range of measurementfrequencies of said sensors are within the range of said antenna meansbut do not overlap.
 12. The acquisition system of claim 8, wherein saidindividual ranges of resonant frequencies correspond to the physicalproperty values to be measured.
 13. The acquisition system of claim 1,wherein said one or more sensors are embedded in material that istransmissive to radio frequency energy.
 14. The acquisition system ofclaim 1, wherein said antenna means is a metallic foil. The acquisitionsystem of claim 1, wherein said antenna means is a thin film depositedon a dielectric membrane.
 15. The acquisition system of claim 1, whereinsaid one or more sensors are fabricated using metal deposition.
 16. Theacquisition system of claim 6, wherein one or more sensors has acapacitor embedded in a conducting material and an inductor placed awayfrom the surface of the conductive material.
 17. The acquisition systemof claim 2, wherein said interrogation means comprises the followingsteps: (a) at the lower limit of a predetermined range, transmitting aradio frequency harmonic for a predetermined length of time from saidantenna; (b) switching said transmission mode of said antenna off; (c)turning the receiving mode of said antenna on; (d) rectifying thereceived response from said sensor to determine its amplitude; (e)storing the amplitude, A_(i)(t), and the frequency, ω_(i)(t), of theresponse; (f) switching the receiving mode off and the transmission modeon; (g) shifting the transmitted radio frequency harmonic by apredetermined amount; (h) transmitting the harmonic for a predeterminedlength of time; (i) switching the transmission mode off; (j) switchingthe receiving mode on; (k) rectifying the received response from saidsensor to determine its amplitude; (l) storing said current amplitude,A_(i), and said frequency, ω_(i); (m) comparing said amplitude, A_(i),to the two previously recorded amplitudes, A_(i-1) and A_(i-2); (n) ifsaid previous amplitude, A_(i-1), is greater than said amplitude, A_(i),and the previous amplitude, A_(i-1), is greater than the amplitude priorto it, A_(i-2), storing said amplitude, A_(i-1), as the amplitudeinflection and the corresponding frequency, ω_(i-1), for the currentfrequency sweep; (o) comparing said amplitudes obtained in step (n) withthe amplitudes of the next subsequent sweep; (p) repeating steps (f)through (l) if an amplitude inflection has not been reached; and (q)once amplitude inflection has been reached, continuing the sweep to saidnext sensor.
 18. The acquisition system of claim 18, wherein the sweeprate for each said sensor is dependent on the rate of change of thephysical state being measured.
 19. The acquisition system of claim 18,wherein said sensors have one or more different resolutions.
 20. Theacquisition system of claim 18, wherein the resolution of one or moresensors is not fixed.
 21. The acquisition system of claim 18, whereindynamic measurements are obtained by comparing variation in one ore moreresponses selected from the group consisting of frequencies, of acurrent sweep with those of prior sweeps.
 22. The acquisition system ofclaim 18, wherein a change in position of a sensor is obtained fromcomparison of amplitude variations of successive sweeps.
 23. Theacquisition system of claim 18, wherein said interrogation meansdetermines amplitude, frequency and bandwidth variation with time. 24.The acquisition system of claim 18, wherein said first sweep determinesall resonant frequencies and corresponding amplitudes.
 25. Theacquisition system of claim 3, wherein said interrogation meanscomprises the following steps: (a) at the lower limit of a predeterminedrange, transmitting a radio frequency harmonic for a predeterminedlength of time from said antenna; (b) turning said transmission antenna;(c) turning said receiving antenna on; (d) rectifying the receivedresponse from said sensor to determine its amplitude; (e) storing theamplitude, A_(i)(t), and the frequency, ω_(i)(t), of the response; (f)turning said receiving antenna off and transmission antenna on; (g)shifting the transmitted radio frequency harmonic by a predeterminedamount; (h) transmitting the harmonic for a predetermined length oftime; (i) turning said transmission antenna off; (j) turning saidreceiving antenna on; (k) rectifying the received response from saidsensor to determine its amplitude;. (l) storing said current amplitude,A_(i), and said frequency, ω_(i); (m)comparing said amplitude, A_(i), tothe two previously recorded amplitudes, A_(i-1) and A_(i-2); (n) if saidprevious amplitude, A_(i-1), is greater than said amplitude, A_(i), andthe previous amplitude, A_(i-1), is greater than the amplitude prior toit, A_(i-2), storing said amplitude, A_(i-1), as the amplitudeinflection and the corresponding frequency, ω_(i-1), for the currentfrequency sweep; (o) comparing said amplitudes obtained in step (n) withthe amplitudes of the next subsequent sweep; (p) repeating steps (f)through (1) if an amplitude inflection has not been reached; and (q)once amplitude inflection has been reached, continuing the sweep to saidnext sensor.
 26. The acquisition system of claim 26, wherein the sweeprate for each said sensor is dependent on the rate of change of thephysical state being measured.
 27. The acquisition system of claim 26,wherein said sensors have one or more different resolutions.
 28. Theacquisition system of claim 26, wherein the resolution of one or moresensors is not fixed.
 29. The acquisition system of claim 26, whereindynamic measurements are obtained by comparing variation in one ore moreresponses selected from the group consisting of frequencies, of acurrent sweep with those of prior sweeps.
 30. The acquisition system ofclaim 26, wherein a change in position of a sensor is obtained fromcomparison of amplitude variations of successive sweeps.
 31. Theacquisition system of claim 26, wherein said interrogation meansdetermines amplitude, frequency and bandwidth variation with time. 32.The acquisition system of claim 26, wherein said first sweep determinesall resonant frequencies and corresponding amplitudes.
 33. Theacquisition system of claim 1, wherein said interrogation meanscomprises: an one antenna for transmitting and receiving a varyingmagnetic field; a microcontroller that places said antenna intotransmission mode and submits a binary code to a frequency synthesizer,said frequency synthesizer concerting said code into a square wave withthe frequency of the wave dependent on said binary code; a high-speedamplifier that amplifies said square wave; a low pass filter thatattenuates all frequencies that are higher than a prescribed frequencyfor application to said antenna for a prescribed number of cycles;applying said low pass filter signal to said antenna for a prescribednumber of cycles; a radio frequency receiving/transmission switch forswitching said antenna to receiving mode; a high speed amplifier thatamplifies the signal from said sensor after it is received from saidantenna; a diode peak detector that rectifies said amplified signal andcreates a DC value proportional to signal amplitude; an op amp thatamplifies said DC voltage from said peak detector; an analog to digitalconverter that converts said signal from said op amp to a digitalsignal; and said microcontroller storing the amplitude of digital signaland the transmission frequency.
 34. The acquisition system of claim 34,comprising separate transmission and receiving antennae.
 35. Theacquisition system of claim 18, wherein data is stored for the entirerange, followed by peak amplitudes being determined for each saidsensor, further wherein said peak amplitudes and correspondingfrequencies are stored for comparisons to subsequent sweeps.
 36. Theacquisition system of claim 26, wherein data is stored for the entirerange, followed by peak amplitudes being determined for each saidsensor, further wherein said peak amplitudes and correspondingfrequencies are stored for comparisons to subsequent sweeps.
 37. Theacquisition system of claim 18, further comprising a data filecorresponding to each said sensor, comprising said sensor type, responsevariation, frequency partition and measurement band for each saidpartition sweep after said resonant is identified on said initial sweepand further comprising a table that correlates response variation to aphysical state, said data files for each said sensor concatenated toform an aggregate file.
 38. The acquisition system of claim 26, furthercomprising a data file corresponding to each said sensor, comprisingsaid sensor type, response variation, frequency partition andmeasurement band for each said partition sweep after said resonant isidentified on said initial sweep and further comprising a table thatcorrelates response variation to a physical state, said data files foreach said sensor concatenated to form an aggregate file.
 39. Theacquisition system of claim 1, wherein one or more said sensors ismetamorphic.
 40. The acquisition system of claim 1, wherein one or moresaid sensors is multifunctional.
 41. The acquisition system of claim 1,wherein at least one said sensor is mounted to a conductive surface,further wherein said sensor has an inductor that has a fixed separationfrom said conductive surface.
 42. The acquisition system of claim 1,wherein at least one said sensor is measuring response of a conductivecavity, further wherein the inductor of said sensor is mounted externalto said cavity at a fixed distance from said cavity wall, the capacitorof said sensor is mounted internal to said cavity, and said antenna ismounted external to said cavity.
 43. The acquisition system of claim 1,wherein multiple sensors measure response of a conductive cavity,further wherein said inductors of said sensors are mounted external tosaid cavity at a fixed distance from said cavity wall, and said antennais mounted internal to said cavity.
 44. The acquisition system of claim1, wherein at least one sensor measures material phase transition. 45.The acquisition system of claim 1, wherein at least one sensor measuresone or more attributes selected from the group consisting of strain,fluid-level, proximity, displacement, shear, torsion, pressure, angularorientation, dielectric level, fluid level, solid particle level,material phase transition, moisture exposure, chemical exposure,stoichemetric changes, wear, bond separation, identification ofconductive materials, relative place orientation, and displacement rate.46. The acquisition system of claim 1, wherein at least one said sensorcomprises one or more interdigital electrodes positioned such that saidelectrodes are parallel to a surface of wear.
 47. The acquisition systemof claim 1, wherein at least one said sensor comprises one or moreinterdigital electroplates.
 48. The acquisition system of claim 48,wherein said sensor further comprises an element positioned between saidelectroplates, said elements selected from the group consisting oftemperature sensitive dielectric, thermomagnetic, and phase transitiondielectric.
 49. The acquisition system of claim 1, wherein at least onesaid sensor comprises two parallel electroplates for displacementmeasurements.
 50. The acquisition system of claim 1, wherein at leastone said sensor comprises a dielectric affixed to a stationaryelectroplate for displacement measurements.
 51. The acquisition systemof claim 1, wherein at least one said sensor comprises n pairs ofparallel electroplates separated by a dielectric medium, for fluid levelmeasurement.
 52. The acquisition system of claim 1, wherein at least onesaid sensor comprises two parallel electroplates separated by adielectric medium, for fluid level measurement.